1 Introduction

Microstrip antennas have been widely used in various applications in wireless communication, due to the advantages of miniaturization, low cost and easy integration with microwave RF circuits compared with other types of antennas (Bakariya et al. 2015; Li et al. 2015, 2016). In recent years, the rapid development of wireless communication has promoted increasingly high demands for multi-band antennas. Multiband antennas can be used to replace multiple antennas mounted on different integrated circuits or wireless handheld devices to make the whole system more simple and compact. It is widely used in WLAN/WIMAX/duplex mode n77 band and long-distance telecommunication broadband transmission. For each working band of a multi-band antenna, the bandwidth, gain and radiation characteristics need to be taken into account to meet the practical requirements.

At present, there are various structures of multiband antennas, such as adopting DGS structure to achieve multi-band (Wang et al. 2013; Pan et al. 2016; Jin et al. 2015), ultra-wideband characteristics (Awadhwal and Bostani 2018; Satam and Nema 2018), using metamaterials (Huang et al. 2015) or meta-surface (Wang et al. 2019) technology to improve the gain and the efficiency, and reconfigurable pattern of an antenna, etc. By introducing slotted structures (Hussain et al. 2020) and multi-branch structure (Chandra et al. 2020), the antenna surface current flow path is effectively increased and thus a compact structure is achieved. Multi-Input Multi-Output (MIMO) antenna technology (Kumar and Nath 2020; Satam and Nema 2020) and loading parasitic elements (Saraswat and Harish 2020) can be used to expand the multi-band bandwidth of the antenna. Hajlaoui (2018) designed a novel three-band microstrip patch antenna. Two parasitic elements and the main radiation patch constitute an electromagnetic band gap (EBG) structure, forming a three-band (S/C/X) resonant mode, but the working bandwidth of each band is narrow. Han et al. (2018) proposed a single-feed multi-band antenna based on parasitic coupling element. By adding T-shaped and folded linear parasitic patch on FR-4 substrate, a compact dimension with three-band characteristics is realized. However, the overall structure of the antenna is three-dimensional, and the maximum relative bandwidth is only 4.1%. A multi-band microstrip antenna is proposed in Boutejdar and Halim (2019) which is composed of a radiation patch and two parasitic open-loop resonators, forming a coupling bridge with the DGS. The obvious disadvantages are narrow bandwidth and high production cost. A design of a gap-coupled rectangular patch antenna with meandered parasitic elements and a circular ring at the feed line is proposed (Patel et al. 2020), and the geometric size is still large though having an increased bandwidth. In References Halim and Boutejdar (2019) and Afrough et al. (2015), the authors reduced the electrical size of the multi-band antenna by adding different degrees of sawtooth parasitic elements. However, the overall structure of the antenna is complex and therefore not practical.

In view of the problems of the multi-band antenna such as complex structure and narrow band in the above literature, this paper designs a three-band microstrip slot grounding antenna characterized by wide bandwidths for each band and compact size. The design is based on a DGS structure which effectively reduces the electrical size of the antenna and inhibits the cross-polarization of the antenna. The upper layer of the antenna adopts a closed parasitic ring in the center of the lower DGS and is coupled by a 50-Ω rectangular microstrip feed line. The electromagnetic wave energy is indirectly fed to the DGS through the parasitic ring, making it easy to adjust the characteristics of each frequency band. A pair of rectangular parasitic patches are set on both sides of the microstrip feed line, which can effectively adjust the impedance characteristics of the three bands to obtain better impedance matching. The proposed antenna realizes broadband transmission in each working band, which is simple, compact, and easy to manufacture.

2 Antenna Structure and Principle Analysis

2.1 Antenna Structure

Figure 1 shows the final optimization model of the three-band broadband antenna based on the parasitic ring and DGS. The 50-Ω microstrip feed line and the parasitic patch for impedance matching are printed on the upper layer of the dielectric substrate. The DGS and the closed parasitic ring components are printed on the lower layer of the dielectric substrate. Epoxy resin (FR-4) with relative dielectric constant εr = 4.4, tangent loss angle tan δ = 0.02 and thickness h = 1.6 mm is selected as the dielectric substrate. Despite the acceptable power loss, the advantage of FR-4 substrate is in its low cost and in that it can effectively adjust the band of antenna to shift toward the low-frequency direction.

Fig. 1
figure 1

Antenna structure model

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2.2 DGS and Parasitic Ring Structure

The concept of DGS was first proposed by Park J.I., a South Korean scholar, at the Asia–Pacific Microwave Conference Proceedings (APMC) in 1999 (Park et al. 1999). The DGS is evolved on the basis of photonic electromagnetic band gap structure. It is constructed by etching regular or irregular, periodic or non-periodic defect areas on the whole ground plane, so as to change the distribution of the current transmission paths from the feed line to the ground. In this paper, an axisymmetric approximate rectangular DGS is used, and two isosceles trapezoid concave regions are introduced on both sides, as shown in Fig. 1b, which increases the flow path of the ground current and reduces the electrical size of the antenna itself. Loading a closed parasitic ring structure in the center of the DGS is equivalent to introducing a LC parallel resonant circuit, which is in parallel with the overall DGS. The equivalent circuit of antenna grounding is shown in Fig. 2.

Fig. 2
figure 2

Antenna grounding equivalent circuit

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The parallel inductances L1 and capacitance C1 shown in Fig. 2 are connected in series to form an equivalent microstrip transmission line of 50 Ω. The DGS and the closed parasitic ring are connected in parallel, and each can be represented as a LC parallel resonant tank. It is obvious that the circuit in Fig. 2 consists of three resonant tanks, a series resonant tank and two parallel resonant tanks, which forms the base structure of a multi-band model. The impedance value Z1 of the single microstrip feed line with open terminal is:

$$Z_{1} = j\left( {\frac{{\omega L_{1} }}{2} - \frac{1}{{\omega C_{1} }}} \right) = j\left( {\frac{{\omega^{2} L_{1} C_{1} - 2}}{{2\omega C_{1} }}} \right)$$
(1)

where ω is the angular frequency.

The LC parallel impedances of the DGS and the closed parasitic ring are Z2 and Z3, respectively:

$$Z_{2} = \frac{1}{{j\omega C_{2} + {1 \mathord{\left/ {\vphantom {1 {j\omega L_{2} }}} \right. \kern-\nulldelimiterspace} {j\omega L_{2} }}}} = j\frac{{\omega L_{2} }}{{1 - \omega^{{_{2} }} L_{2} C_{2} }}$$
(2)
$$Z_{3} = \frac{1}{{j\omega C_{3} + {1 \mathord{\left/ {\vphantom {1 {j\omega L_{3} }}} \right. \kern-\nulldelimiterspace} {j\omega L_{3} }}}} = j\frac{{\omega L_{3} }}{{1 - \omega^{{_{2} }} L{}_{3}C_{3} }}$$
(3)

When Z1 + (Z2//Z3) = 0, the antenna is under the resonant state:

$$j\left[ {\frac{{\omega L_{2} L_{3} }}{{L_{2} \left( {1 - \omega^{2} L_{3} C_{3} } \right) + L_{3} \left( {1 - \omega^{2} L_{2} C_{2} } \right)}} + \frac{{\omega^{2} L_{1} C_{1} - 2}}{{2\omega C_{1} }}} \right] = 0$$
(4)

The imaginary part in formula (4) can be converted to:

$$aw^{4} + b\omega^{2} + c = 0$$
(5)

where the expressions of the coefficients a, b and c are:

$$a = L_{1} L_{2} L_{3} C_{1} \left( {C_{2} + C_{3} } \right);$$
$$b = 2L_{2} L_{3} \left( {C_{1} - C_{2} - C_{3} } \right) - L_{1} \left( {L_{2} + L_{3} } \right)C_{1} ;$$
$$c = 2\left( {L_{2} + L_{3} } \right).$$

To solve the angular frequency ω, we can apply the factorization method to formula (5):

$$\begin{aligned} & \left( {\omega + \sqrt {{{\left( { - b + \sqrt \Delta } \right)} \mathord{\left/ {\vphantom {{\left( { - b + \sqrt \Delta } \right)} {2a}}} \right. \kern-\nulldelimiterspace} {2a}}} } \right)\left( {\omega - \sqrt {{{\left( { - b + \sqrt \Delta } \right)} \mathord{\left/ {\vphantom {{\left( { - b + \sqrt \Delta } \right)} {2a}}} \right. \kern-\nulldelimiterspace} {2a}}} } \right) \\ & \left( {\omega + \sqrt {{{\left( { - b - \sqrt \Delta } \right)} \mathord{\left/ {\vphantom {{\left( { - b - \sqrt \Delta } \right)} {2a}}} \right. \kern-\nulldelimiterspace} {2a}}} } \right)\left( {\omega - \sqrt {{{\left( { - b - \sqrt \Delta } \right)} \mathord{\left/ {\vphantom {{\left( { - b - \sqrt \Delta } \right)} {2a}}} \right. \kern-\nulldelimiterspace} {2a}}} } \right) \\ & \quad = 0 \\ \end{aligned}$$
(6)
$$\Delta = b^{2} - 4ac$$
(7)

where b needs to be negative; otherwise, the angular frequency ω would have complex roots. So the resonant frequency fs is obtained:

$$f_{s} = {\omega \mathord{\left/ {\vphantom {\omega {2\pi }}} \right. \kern-\nulldelimiterspace} {2\pi }}$$
(8)

From Eqs. (6) and (8), it can be seen that the antenna structure has four resonant frequencies. The return loss curve of the DGS structure loaded with closed parasitic ring is shown in Fig. 3. Since the return loss at the center frequency of 9.14 GHz is greater than -10 dB, the corresponding resonant frequency here can be ignored. The closed parasitic ring plays a vital role in producing the required resonant points. Due to the existence of the closed parasitic ring, the current flow direction on the ring surface changes with different resonant frequencies, as shown in Fig. 4. After removing the parasitic ring in the center of DGS, the antenna can only realize data transmission in single-frequency band as shown in Fig. 3.

Fig. 3
figure 3

Comparison curve of return loss between loaded parasitic ring and unloaded parasitic ring

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Fig. 4
figure 4

Current flow path on the surface of parasitic ring and current distribution in DGS

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2.3 Impedance Matching Network Analysis of Feed Structure

A microstrip transmission feed line with an open terminal, combined with two symmetric rectangular parasitic patches on both sides of it, forms a feeding unit which effectively adjusts the impedance matching of the antenna. According to the current distribution on the microstrip feed line, the input impedance at the central frequency point of each band can be optimized by adjusting the size and position of the parasitic patch. The initial length Lg of the patch is set to 0.1λ, and the width Wg is set to 0.1λ. The length Lm of the feed line is set to 0.35λ, and its width Wm is set to 0.05λ. The interval of the parasitic patch is 0.08λ, where λ is the wavelength of the antenna. In different operating frequency bands, when the excitation current flows on the feed line, the surface current distribution is different which in turn causes the current distributions coupled to the rectangular patch on both sides to be different. For the frequency out-of-band, the coupling currents generated on the parasitic patches are out of phase or even inverse. As a result, the radiation is suppressed. This feeding mode has a simple structure and is easy to design, adjust and optimize.

The distance between the parasitic patch center and the feed line terminal is λ/4, as shown in Fig. 5a. The corresponding feed line impedance is short circuit, so the feed structure as a whole can be equivalent to a LC series resonant circuit, as shown in Fig. 5b. The current on the microstrip line in the area near to the patch center is the strongest, and the corresponding magnetic field distribution reaches the maximum. Therefore, a strong magnetic coupling occurs between the parasitic patch and the microstrip transmission line, which is represented as the series inductance La in the equivalent circuit. The gap between the two parasitic patches can be equivalent to the series capacitor Ca.

Fig. 5
figure 5

Feed structure and equivalent circuit

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The distance n2 between the symmetric rectangular parasitic patch and the microstrip feed line can be adjusted to effectively match the input impedance. The input impedance at various values of spacing n2 is matched and optimized by HFSS software, as shown in Fig. 6. When the distance n2 increases, the electromagnetic energy coupled by the feed line is reduced and thus the series capacitance Ca decreases. In the middle frequency band, the real impedance increases gradually, and the imaginary impedance decreases. At the high-frequency band, the impedance changes of the real part and the imaginary part are consistent with each other, showing a decreasing trend. After some simulations, the distance of n2 is finally determined as 0.5 mm; as a result, the corresponding optimized input impedance at the center frequency of the three working bands is (49.9439 + j0.1170) Ω, (54.3387 + j3.7789) Ω, (51.6748 − j5.3941) Ω, respectively.

Fig. 6
figure 6

Impedance matching optimization curve

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3 Antenna Simulation and Actual Measurement Analysis

Based on the principle analysis of the DGS above, the performance optimization of some structural parameters of the antenna will be carried out in this section. The influence of these parameters on the working band and impedance bandwidth of the antenna will be further analyzed. The final antenna structural parameters determined by simulations are shown in Table 1.

Table 1 Antenna final size parameter
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3.1 Antenna Parameter Optimization

On the basis of the antenna parameters given in Table 1, the outer ring width Ws, length Ls and ring width g of the antenna connected to the ground parasitic element are optimized and analyzed by the parameter scanning function of HFSS. The parameter scanning of the trapezoidal depression depth m4 on both sides of the DGS is carried out. It can be seen from Fig. 7a that the outer ring width Ws of the closed parasitic ring can effectively affect the low-frequency band and medium-frequency band of the antenna. When the value of Ws is increased from 24 to 25 mm in a step of 0.5 mm, the resonant frequency fs of the low-frequency band begins to move to the right, and the relative bandwidth is improved. The change of Ws affects the medium-frequency band more obviously. As the outer ring width increases, both the resonant frequency fs and the cutoff frequency fc of the medium-frequency band move to the left. Taking into consideration the required isolation between the low-frequency band, the middle-frequency band and the overall effect of the return loss curve, the Ws is set to 24.5 mm.

Fig. 7
figure 7

The influence of the changes of antenna parameters on the return loss

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The outer ring length Ls of the parasitic element on the grounding layer has effects on the frequency characteristics of the antenna in the middle and high bands, as shown in Fig. 7b. As the outer ring length Ls increases, the mid-band and high band shift to the right and the left, respectively, causing the associated reflection coefficient and bandwidth to deteriorate. The change of Ls value has no significant effect on the low-frequency section. The width of the parasitic ring g only affects the return loss of the high-frequency band. With the increase of the ring width g, the center frequency fs and the left end cutoff frequency fc shift to the left, as shown in Fig. 7c. The reason is that in the equivalent LC parallel resonant circuit with closed parasitic ring, the equivalent inductance L3 gradually decreases with the increase of g, while the equivalent capacitance C3 increases slowly. It can be seen that the parallel equivalent inductance L1 and series equivalent capacitance C1 of the open-circuit microstrip transmission line are also reduced.

The isosceles trapezoid depression depth in the DGS also has a significant effect on the characteristics of the high-frequency band. When the value of m4 increases with the step length of 1 mm, the resonant frequency of the high-frequency band tends to shift to the right, as shown in Fig. 7d. However, the cutoff frequency fc of this band does not shift at this time; thus, it does not affect the impedance bandwidth performance. The trapezoidal concave structure in DGS can increase the length of the flow path of surface currents in the grounding layer, which effectively reduces the electrical size of the antenna to a certain extent. It can also be seen that if the depression depth of the isosceles trapezoid becomes larger, the inductance L2 in the equivalent circuit increases gradually, and the capacitance C2 decreases continuously. At the same time, with the increase of m4 value, the radiation efficiency in the working bands will also be affected to various degrees. As shown in Fig. 8, in the three operating frequency bands, the radiation efficiency is about 20%, 30% and 40%, respectively. When m4 increases from 3 to 4 mm, the radiation efficiency decreases by about 4%, with the flatness being still good in each operating frequency band. It is shown obviously that the efficiency decreases with the increasing depression depth. In other words, there is a trade-off between miniaturization and radiation efficiency. Therefore, when using the defective structure to realize the miniaturization, it is necessary to optimize m4 to ensure that the overall performance is not greatly affected. According to the simulation and optimization results of the reflection coefficient and radiation efficiency, it is finally determined that the value of m4 is set to 3 mm. This optimized parameter assures that the bandwidth of the antenna in the high-frequency band can reach 1.07 GHz (7–8.07 GHz), and the relative bandwidth is about 14.2%, which meets the requirements of broadband antenna. Meanwhile, the reflection coefficient at 7.62 GHz is increased to − 23.18 dB.

Fig. 8
figure 8

The variation curve of radiation efficiency with m4 value

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3.2 Performance Comparison Between Simulation and Measurement

In order to verify the practical performance of the designed three-band broadband antenna based on the DGS and loaded parasitic elements, the antenna prototype was fabricated according to the optimized dimension parameters of the antenna in Table 1, as shown in Fig. 9. The vector network analyzer is used to measure the return loss of the antenna. The simulation results are compared with the measured results as shown in Fig. 10. The return loss is less than -10 dB in the frequency range in 2.10–2.67 GHz, 3.35–4.15 GHz and 7–8.07 GHz. The corresponding relative bandwidths are 23.9%, 21.3% and 14.2%, respectively, which meets the requirements of a broadband microstrip antenna. The measured results are in good agreement with the simulation results. The resonant frequency points in the middle and low bands slightly shift to the right, which may be caused by factors such as antenna fabrication tolerance or test environment interference. The three working frequency bands of the antenna are all broadband, which means the frequency offset within a certain range has little effect on the practical application.

Fig. 9
figure 9

Antenna prototype

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Fig. 10
figure 10

Comparison of measured S11 and simulated S11

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Figure 11 shows the simulated gain of the designed antenna in comparison with the measured results. In the low-frequency band of 2.1–2.67 GHz, the antenna gain first increases and then decreases. The measured result shows that the maximum gain is 3.08 dB at 2.45 GHz. In the band of 3.35–4.15 GHz, with the increasing frequency the antenna gain maintains a slight increasing, and the peak gain of 4.15 GHz reaches 4 dB in this band. In the high-frequency band, the peak gain of the antenna fluctuates greatly compared with that of the low-frequency band. In the whole range from 1 to 10 GHz, the gain varies from 5.25 to 6.15 dB with the maximum gain of 6.15 dB at 7 GHz. The measured and simulated curves are in good agreement, in spite of a negligible deviation that the peak gain in each working frequency band is about 0.5 dB smaller than the simulation results.

Fig. 11
figure 11

Comparison curve between simulated and measured gain

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At the resonant frequencies of 2.26 GHz, 3.81 GHz and 7.62 GHz, the radiation patterns of the antenna with a closed parasitic ring and the one without it are shown in Fig. 12. E-plane and H-plane represent the plane of electric field vector and magnetic field vector, respectively. In the E-plane, the radiation patterns at the three resonant frequencies are approximately ‘8.’ Though there are different degrees of distortion at the high-frequency point of 7.62 GHz, the half-power lobe width still exceeds 60°. In the H plane, at the resonant frequencies of 2.26 GHz and 3.81 GHz, an approximate symmetric omni-directional radiation pattern is presented and the radiation performance is relatively stable, with the lobe width reaching 100°. By comparison, the peak gain of the loaded parasitic ring is slightly higher than that of the unloaded parasitic element in both E and H planes.

Fig. 12
figure 12

Simulation results of E-plane and H-plane antenna radiation patterns

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4 Antenna Performance Comparison

Table 2 is the performance comparison between the designed antenna and the existing antenna in other references. It shows that the relative bandwidth of the three bands of the antenna designed in this paper is relatively balanced and meets the requirements of the broadband characteristics of a microstrip antenna. Compared with the multi-band antenna proposed in references Hajlaoui (2018) and Han et al. (2018), the antenna designed in this paper has advantages in the working bandwidth of each frequency band. The antennas designed in references Boutejdar and Halim (2019) and Halim and Boutejdar (2019) used Rogers RO3010 material as the substrate, and the prototype is more costly and has a narrow relative bandwidth. In reference Patel et al. (2020), the antenna has five bands, of which 1.83–3.81 band shows ultra-wideband characteristics, but the relative bandwidth distribution of each band is not balanced, and the geometric size and radiation gain are less than those designed in this paper. The antenna structure in reference Afrough et al. (2015) is complex and the profile is high. The antenna model designed in this paper has the advantages of multi-frequency broadband, compact structure and low cost, and well meets the practical application in various occasions.

Table 2 Performance comparison between designed antenna and other antennas
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5 Conclusion

In this paper, a new type of DGS microstrip antenna covering three frequency bands is designed. In addition, the parasitic elements are combined to improve the performances of frequency characteristics and the impedance matching. An equivalent circuit is developed to analyze the principle of the proposed antenna. The final optimized working bands of the antenna are in range of 2.10–2.67 GHz, 3.35–4.15 GHz and 7–8.07 GHz, with the relative bandwidths 23.9%, 21.3% and 14.2%, respectively. There is a novel attempt to load a closed parasitic ring in the center of the defect ground. In general, the antenna has the advantages of compact and simple in structure, multi-frequency broadband and good radiation direction, which meets the applications of WLAN/WIMAX/duplex mode n77 band/long-distance telecommunication broadband transmission.