1 Introduction

It is expected that by the end of 2021, the next generation of telecommunications networks (fifth generation or 5G) will hit the market and will continue to expand globally. 5G is expected to create a large IoT (Internet of Things) ecosystem in addition to data rate enhancement, in which networks can meet the connectivity needs and demands of billions of linked devices, coping with speed, latency and costs as necessary. For 5G networks (millimeter waves between 30 and 300 GHz), using lower frequencies is why 5G can be faster. By transmitting information through millimeter waves [1], 5G mobile communications systems have a major effect on modern technology. As an example of progressive improvements in the network interface [2], compared with current cellular infrastructure, this represents a major move forward. 28 GHz and 38 GHz [3] are the major bands intended for the future 5G mobile wireless networks and to fulfill bandwidth requirements. As regards the analysis of the under-used mm-wave frequency range, several works on the propagation study of cellular mm-wave in densely populated environments [4,5,6,7,8] using various shapes of smart antennas have been published for future broadband wireless communication networks. For instance, T. Recently, Rappaport et al. published detailed propagation observation campaigns at 28 GHz and 38 GHz bands to provide insight into their indoor and environmental applicability in Austin and New York City [9]. In addition, narrowband and wideband observations have been carried out under various indoor and outdoor conditions to allow precise predictive models to be generated for 60 GHz radio channels [10]. But there are limitations in the antenna designs used in this literature in terms of cost, size and complex circuitry. Use of the reconfigurable antennas, on the other hand, is another interesting and exciting approach to the new developments in wireless communications networks [11]. This community of antennas helps to reconfigure not only the bandwidth, but also the radiation and polarization patterns. Once it provides agile frequency, the reconfigurable antenna offers flexibility in architecture, using specified operating system and cognitive radios to deal with multi-service, multi-standard and multi-band operations that can be extended and reconfigured. Furthermore, using reconfiguration, an efficient utilization range as well as power consumption is achieved. The Adaptive and Cognitive Radio over Fiber (ACROF) concept [12] is recently proposed, focusing on the combination of optically controlled reconfigurable antennas and ROF systems [13]. A high-capacity optical back-haul with a wireless, energy and spectrum-efficient remote access module is feasible using this new approach, and fresh optical advantages are also being used. A lightweight, reconfigurable hybrid antenna is presented by the authors of [14], which has two connectors used for switching the surface current distribution, frequency and radiation pattern of the antenna. In the different wireless communication bands with steerable beam pattern designed on graphene pads, the reconfigurable 5G to 4G operating band for radio frequency energy harvesting (RF-EH) is suggested [15]. A reconfigurable microstrip ultra-wide band (UWB) antenna, where frequency and pattern switching operating within the 3 GHz to 10 GHz frequency range is presented by the authors of [16]. Using PIN diodes or NMOS transistor as switching instruments, reconfigurability of the frequency and pattern is achieved [16]. A basic pattern reconfigurable printed Inverted-F antenna composed of one main radiator and one parasitic component is proposed for WLAN applications with low profile monopole antenna [17]. The proposed antenna, using only one PIN diode, can change the pattern and the polarization over the entire 205 MHz range.

This research work provides a state-of-the-art method for introducing hybrid reconfigurable features of frequency and radiation pattern in a microstrip patch antenna. Depending on the state of the lumped switch S3, the proposed antenna operates at the desired frequency bands of 28 GHz and38GHz, while the other two switches, S1 and S2 are installed in the ground to create reconfigurability of the radiation pattern.

2 Statistics of Reconfigurable Antenna Patch

The proposed reconfigurable patch antenna model typically consists of a patch, often manufactured on a substrate and ground with Lumped parameter. Five geometric parameters (\(L,W,g,L_{d} \,{\text{and}}\,{\text{ }}LP_{k}\)) (Eq. 1.) adjusted for kth number of lumped parameter [18, 19] are included in the architecture of the HRA with inset feed and consist of three layers including patch substrate and parasitic stubs. Whereas both the parasitic stubs and the patch of the antenna can be connected via \(LP_{k}\) (kth number of Lumped Parameter, used for switching purpose).

$$ \left\{ \begin{gathered} \Phi (W,L,y_{0} ,g,W_{0} ,.LP_{k} ) = [\delta B,\delta A_{e} ,\delta G].{\text{ }}W^{T} \hfill \\ Obj.Fun = \min \Phi (W,L,y_{0} ,g,LP_{k} ) \hfill \\ \end{gathered} \right. $$
(1)

W, width of patch; L, length of patch; y0, any required slot length; g, notch width; W0, feed with of patch

$$ W = \frac{c}{{2f_{0} \sqrt {\frac{{\varepsilon _{{{\text{eff}}}} + 1}}{2}} }} $$
(2)

\(\varepsilon _{{{\text{eff}}}}\), effective dielectric constant; f0, resonant frequency.

Because of the fringing fields and effective dielectric constant Eq. (3), length of the patch is calculated from Eq. (5);

$$ \varepsilon _{{{\text{eff}}}} = \frac{{\varepsilon _{r} + 1}}{2} + \frac{{\varepsilon _{r} - 1}}{2}\left( {\frac{1}{{\sqrt {1 + 12\left( {\frac{h}{W}} \right)} }}} \right) $$
(3)

\(\varepsilon _{r}\), dielectric constant of material; h, separation of the trace from the ground (thickness).

Patch Length:

$$ L = L_{{{\text{eff}}}} - 2\Delta L $$
(4)

Effective Length:

$$ L_{{{\text{eff}}}} = \frac{c}{{2f_{0} \sqrt {\varepsilon _{{{\text{eff}}}} } }} $$
(5)
$$ \Delta L = 0.412h\left[ {\frac{{\left( {\varepsilon _{{{\text{eff}}}} + 0.3} \right)\left( {\frac{W}{h} + 0.264} \right)}}{{\left( {\varepsilon _{{{\text{eff}}}} - 0.258} \right)\left( {\frac{W}{h} + 0.8} \right)}}} \right] $$
(6)

where δF, δB, δAe and δRL shows the deviation between the performance of the simulated reconfigurable antenna and the performance of the measured reconfigurable antenna (Fig. 1). \(LP_{k}\) (kth number of Lumped Parameters) is actually the forward biased (ON State) and Reversed biased (OFF State) characteristics of a switch (PIN Diode, \(D_{k}\)) in this case.

Fig. 1
figure 1

General architecture of reconfigurable antenna

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2.1 ON State (Forward Biased)

When a PIN diode is forward biased, holes and electrons are injected from the P and N regions into the I-region (as shown in Fig. 3a–c). These charges do not recombine immediately, Instead, a finite quantity of charge always remained stored and results in a lowering of the resistivity of the I-region (Eqs. 7–10).

$$ Q = I_{F} \tau $$
(7)

Quality of stored charge depends on \(\tau\), Recommendation or carrier life time; IF Forward biased current.

The resistance of the I-region under forward bias, RS is inversely proportional to Q and may be expressed as.

Where,

$$ R_{S} = \frac{{W_{I} ^{2} }}{{(\mu _{N} + \mu _{P} )Q}} $$
(8)

WI, I-region width; \(\mu _{N}\), electron mobility; \(\mu _{p}\), hole mobility.

From Eqs. 2 and 3, RS as inverse function of current is

$$ R_{S} = \frac{{W_{I} ^{2} }}{{(\mu _{N} + \mu _{P} )\tau I_{F} }} $$
(9)

Equation 4 is valid over an extremely high frequency range PIN diodes are used in a circuit.

Hence,

$$ D_{k} ({\text{ON}}\,{\text{state}}) = R_{{Sk}} = \frac{{W_{{I(k)}} ^{2} }}{{(\mu _{{N_{k} }} + \mu _{{Pk}} )\tau _{k} I_{{F_{k} }} }} $$
(10)

where k is from 1 ↔ Nth.

2.2 OFF State (Reversed Biased)

At high RF frequencies when a PIN diode is reverse biased, it appears as a parallel plate capacitor, essentially independent of reverse voltage, having a value of (Eqs. 11–13).

$$ C_{T} = \frac{{\varepsilon A}}{W} $$
(11)

\(\varepsilon\), silicon dielectric constant; A, junction area; W, I-regiont thickness.

The lowest frequencies at which this effect begins to predominate is related to the dielectric relaxation frequency of the I-region, \(f_{\tau }\), which may be computed as

$$ f_{\tau } = \frac{I}{{2\pi \rho \varepsilon }} $$
(12)

where, \(\rho\), I-region resistivity.

Hence,

$$ D_{k} ({\text{OFF}}\,{\text{state}}) = C_{{T_{k} }} = \frac{{\varepsilon _{k} A_{k} }}{{W_{k} }} $$
(13)

where k from 1 ↔ Nth.

3 Proposed Reconfigurable Antenna Architecture

A 28/38 GHz reconfigurable antenna is proposed in this section (Fig. 2a) and is fabricated depicted in Fig. 2c, d. The tiny geometry of the antenna is \(23\,{\text{mm}}~ \times ~25\,{\text{mm~}} \times ~0.506\,{\text{mm}}\) is excited through a \(\frac{\lambda }{4}\) transformer with a width of \(w_{1} = ~1.1\,{\text{mm}}\) to the 50 Ω microstrip line with a width of \(w_{7} = ~2.5\,{\text{mm}}\). The antenna dimensions are tabulated in Table 1.

Fig. 2
figure 2

Proposed antenna structure a proposed antenna b fabricated front and back view of prototype in anechoic chamber c fabricated prototype front view d fabricated prototype back view

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Table 1 Different parameter with their values of the proposed antenna
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To calculate the patch width (W), effective permittivity (\(\varepsilon _{{{\text{eff}}}}\)), patch length (L), effective length (\(L_{{{\text{eff}}}}\)), and additional patch length (ΔL) Eqs. (2) to (6) are used [18]. The following procedure was adopted for a simple patch of 38 GHz, which was further modified by introducing stubs and lumped parameters.

Using Eqs. 2–6, Calculation for 38 GHz Frequency.

From Eq. 2

$$ W = \frac{{3 \times 10^{8} }}{{2\left( {38 \times 10^{9} } \right)\sqrt {\frac{{2.33 + 1}}{2}} }}{\text{ }} $$
$$ W = 3\,{\text{mm}} $$
(14)

By placing Eq. 14 in 3, we get

$$ \varepsilon _{{{\text{eff}}}} = \frac{{2.33 + 1}}{2} + \frac{{2.33 - 1}}{2}\left( {\frac{1}{{\sqrt {1 + 12} \left( {\frac{{0.506}}{3}} \right)}}} \right) $$
$$ \varepsilon _{{{\text{eff}}}} = 2 $$
(15)

Similarly,

$$ L_{{{\text{eff}}}} = \frac{{3 \times 10^{8} }}{{2\left( {38 \times 10^{9} } \right)\sqrt 2 }} $$
$$ L_{{{\text{eff}}}} = 2.8\,{\text{mm}} $$
(16)

From Eq. 6

$$ \Delta L = 0.412\left( {0.506} \right)\left[ {\frac{{\left( {2 + 0.3} \right)\left( {\frac{3}{{0.506}} + 0.264} \right)}}{{\left( {2 - 0.258} \right)\left( {\frac{3}{{0.506}} + 0.8} \right)}}} \right] $$
$$ \Delta L = 0.4\,{\text{mm}} $$
(17)

Therefore, from Eqs. 16 and 17, we get

$$ L = 2.8 - 2\left( {0.4} \right) $$
$$ L = 2\,{\text{mm}} $$
(18)

Similar procedure was also adopted for 28 GHz frequency. ON/OFF states are operated by using the PIN diode as a switch. An inductor in series with a resistor is used for the ON state, while in parallel with a capacitor for the OFF state, with reference to the compatibility list of PIN diode SPDT switch (Fig. 3).

Fig. 3
figure 3

PIN diode equivalent circuit a forward bias and revers bias ON/OFF states b equivalent simulator model c cross section basic PIN diode

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4 Results and Discussion

The reflection coefficients for different combinations of bias voltages covering the configured frequency band are shown in this section. Return loss (RL) of the proposed antenna is reconfigured under ON and OFF states of S3. The proposed architecture was selected according to the standard requirements of 5G cellular communication. The circuit board was fabricated using a procedure of optic-radiation of photolithography, which processed the photoresist layer to paste the mask onto silicon sheets. The proposed antenna was designed in a high frequency structure simulator Electromagnetics Suit, Ansys, Canonsburg, PA, USA, HFSS (High Frequency Structure Simulator) 19.2 3D electromagnetic field for the radio frequency and wireless design, and was measured using a ZVA 40 GHz vector network analyzer.

The reflection coefficients for different combinations of bias voltages covering the tuning frequency band are shown. The return loss S11 (dB) response of the proposed antenna has been made reconfigurable under ON and OFF states of the Switch 3 and can be viewed in Fig. 4a, b. Switch 3 can be seen clearly, which causes the frequency reconfiguration, between 28 and 38 GHz. Effects of various states of switches as given in Table 2, are discussed. During Case1 (from Mode1-Mode3) S3 is kept ON (connecting P1 & P2), for 28 GHz which results in an RL of app. -32.3 dB as shown in Fig. 4a. Similarly, when P1 & P2 are disconnected (Case2) by deactivation of S3 (OFF position), results in a RL of app. − 42.1 dB at resonant frequency of 38 GHz, as shown in Fig. 4b. At both the cases, different modes of switches are summarized in Table 2.

Fig. 4
figure 4

Reflection coefficient at a S3 is OFF b S3 is ON c VSWR at S3 ON and d VSWR at S3 OFF

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Table 2 Different cases of switches
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Figure 4c, d shows the simulated and measured VSWR for every case (from Table 2). The results show that, the impedance bands were 1.51 GHz and 1.57 GHz for 28 GHz and 38 GHz (< 2.5 standards, respectively).

The size and configuration of the ground plane is an important factor in the assessment of the impedance and radiation properties of the antenna due to time variations on the ground plane. The addition of H-shaped stub on the surface has very little effect on impedance, but there is a huge impact on the proposed antenna radiations. In all switching conditions, there is a particular current distribution that contributes to the specific beam steering trend.

Figure 5 shows radiation pattern results from Table 3 (Mode 1 to 6). It illustrates that at Case 1, when S3 is ON, the antenna operates with a resonant frequency of 28 GHz (it is unconcerned with 38 GHz). At this stage, the radiation beam direction of the antenna is such as, it radiates between 90° to 270° at Mode 1 (S1 and S2 both are at ON position) (Fig. 5a), 0° to 180° at Mode 2 (S1 is at ON position and S2 is at OFF position) (Fig. 5b), and 180° to 360° at Mode 3 (S1 is at OFF position and S2 is at ON position) (Fig. 5c). Similarly, during Case 2, when S3 is kept OFF and proposed antenna resonates at 38 GHz, the beam direction is rotated as, it radiates between 90° to 270° at Mode 4 (S1 and S2 both are at ON position) (Fig. 5d), 0° to 180° at Mode 5 (S1 is at ON position and S2 is at OFF position) (Fig. 5e), and 180° to 360° at Mode 6 (S1 is at OFF position and S2 is at ON position) (Fig. 5f).

Fig. 5
figure 5

Radiation pattern at a Mode1 b Mode2 c Mode3 d Mode4 e Mode5 f Mode6

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Table 3 Gain and efficiency for mode1 and mode4
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The changing positions of the pattern (anti-clockwise) cause the surface current density to vary, due to the connecting parasitic stubs H. Depending on the number of stubs, we can increase/decrease the steering angle (45° shift is used in this paper). Gain and efficiency with optimal values have been shown in Table 3. The acceptable level of VSWR for wireless application should be less than 2 dB as shown in Fig. 4c, d.

5 Conclusion

A novel lightweight hybrid reconfigurable antenna is modeled and experimentally tested in this research work for pattern and frequency switching between 28 and 38 GHz resonant frequencies. RTDuroid5880 with \(\varepsilon _{r} = 2.33\), \(\tan \delta = 0.0009\) and thickness (h) of \(0.506\,{\text{mm}}\) is used as a substrate material and the antenna is excited through an inset feed. The designed antenna can be utilized for advanced 5G mobile communication applications due to its competitive features. Depending on the states of S1, S2 & S3 frequency and radiation pattern switching is achieved. Main beam of the antenna is circulated through S1 and S2 by connecting parasitic stubs H1 and H2 and the frequency is configured by S3 between 28 and 38 GHz (connecting patch P1 with P2). The proposed antenna operates at 28 GHz @RL ≤ − 32.3 dB and when S3 is in-active (at OFF state), it results in RL ≤ − 42.1 dB at 38 GHz (published literature comparison is presented in Table 4).

Table 4 Performance comparison with previously published work
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