1 Introduction

Ultra-wideband (UWB) devices use narrow or short-duration pulses, making massive or wideband transmission data transfer possible. UWB innovation holds a good agreement for an enormous range of new applications. It accepts unique advantages to organizations and buyers. It can work by using the capacity that involves existing radio services without interference, allowing the proficient use of range assets [1]. The compact antenna for the UWB range communication system has achieved convenience attention in industrial and academic communities. Due to the high data transmission and reception processing rate, simple hardware configuration, low power consumption, and low-cost design, the dielectric resonator antenna (DRA) is preferred. UWB antennas are essential in many systems, including electronic warfare, radar, medical imaging, and wireless communication. Several DRA has been studied to date, which vary structural designs [2, 3]. DRA has attained agreeable consideration for designing the UWB antenna due to its remarkable features [4,5,6,7] indicated by the previous work of Majeed on UWB from 6.6 GHz to 11.3 GHz [8].

The main noticeable features of dielectric resonator (DR) antennas (DRAs) have low metallic losses, wide bandwidth, lightweight, and high radiation efficiency [9, 10]. Short-range communication faces a rich scattering environment, which gives problems like fading due to the multipath and power loss due to disintegrating conditions or mismatch of the transmitter and receiver antenna. DRA has various techniques to achieve CP, including the modification of the shape of DR [11,12,13], metallic strips [14], Cross-slot-coupled [15], and single/dual feeding structure [16, 17]. With a 3 dB axial ratio (AR) bandwidth, an ultra-wideband DRA is used as circular polarised (CP) for wireless applications [18]. In some methods, the CP depends on the structure of the DR. Therefore, the DRA shape is modified to be as asymmetric (width and height) structure for the broadband CP response in the UWB region.

A compact asymmetrical structured DRA with an F-shaped metallic strip in the ground plane is used to achieve wider AR bandwidth (3-dB) for the UWB operations. The proposed antenna structure is designed, analyzed, and optimized using CST microwave studio software. The reflection coefficient of the proposed antenna is measured using a KEYSIGHT (E5063A) vector network analyzer, and the far-field results of the antenna are obtained by the anechoic chamber. The present work is organized as follows: Section 2 presents the proposed antenna geometry and configuration. Section 3 describes the parametric optimization to achieve the desired result. Section 4 presents simulated and measured results. The conclusion for the proposed antenna is presented in section 5.

2 Antenna geometry and configuration

The geometry of the circularly polarized asymmetrical DRA is shown in figure 1. It consists of two rectangular blocks, having the same permittivity εDR = 9.8. These blocks are called dielectric resonators (DRs), first and second block height, length, width are tdr1, tdr2, ldr1, ldr2, wdr1, wdr2, respectively, as shown in figures 1 (a) and (b). The complete design is printed on a low-cost FR4 substrate with a permittivity εs = 4.4 with dimensions (Ls × Ws × ts) as 25 × 25 ×0.8 mm3. On top of this substrate, a microstrip feedline is printed to excite the DR radiator with 50 Ω input impedance. The feeding network ensures proper impedance matching in a wide range of frequencies. The microstrip feed-line have two parts lf1 × wf1 (50 Ω) and lf2 (λ/4) × wf2, which are added to each other. The metallic F-shaped strip added with a partial ground plane is printed on the bottom side of the substrate. The partial ground plane has dimensions Lg × Wg, and F-shape metallic strip dimensions are lt1, lt2, lt3, st1, st2, as shown in figures 1 (a and b). The metallic strips are used in DRA to improve overall gain and AR bandwidth [15, 16]. The perspective and side view of the antenna are shown in figures 1(c)–(d). All design parameters with optimized values are given in Table 1.

Figure 1
figure 1

Compact CPDRA (a) Front view, (b) Back view, (c) Perspective and (d) Side view.

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Table 1 Optimized dimensions of the proposed antenna.
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2.1 Antenna operation

The complete evolution of the final antenna (Ant. 5) is shown in five different stages as Ant.1, Ant.2, Ant.3, Ant.4, and Ant.5, respectively, shown in figure 2. All antennas except Ant. 1 can be employed for wideband applications, as shown in figure 3(a). In the second stage (Ant. 2), DR and the partial ground plane achieve the wide impedance bandwidth (70.3%) but not reported 3-dB AR bandwidth. In the third stage (Ant. 3), an F-shaped metallic strip is added with the partial ground, which enhanced the peak gain from 3.7 dB to 5.3 dB, as observed in figure 3(b). In the fourth stage (Ant. 4), a second block (Red) is placed above the initial block (the pink), which gave two different sections with height 4 mm (DR1) and 8 mm (DR2) to achieve the 3-AR bandwidth of 17.4%. In the final stage (Ant. 5), the width of the DR1 is increased to achieve 3-dB AR bandwidth of 63.8%, as observed in figure 3(c). Table 2 shows the performance of the designed antenna by five different stages.

Figure 2
figure 2

Step evolution of the proposed antenna.

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Figure 3
figure 3

Frequency response concerning (a) Reflection coefficient, (b) Gain and (c) Axial Ratio.

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Table 2 Performance of all five antennas.
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2.2 Widen bandwidth mechanism

The broad bandwidth mechanism is discussed in this section using the current distribution on the metallic surface and the DR perturbation theory. Firstly, the current distribution of the antenna is presented in figures 4a–d. The current rotation is observed from the surface current distribution on the full, with DGS and partial ground plane, as shown in figure 4. Also, it can be observed that the decoupling structure provided the maximum surface current on the edges of the decoupling slot with different frequencies. The current distribution is shown in figures 4(a)–(b) at 7.2 GHz with single resonance path. Figures 4c–d offers the current cancellation having two resonance paths at 9 GHz. As compare to full ground, the modified ground is reported good impedance bandwidth, as shown in figures 4(b) and (d).

Figure 4
figure 4

Surface current distribution on the full ground and modified ground; a–b 7.2 GHz, c–d 9 GHz.

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Secondly, perturbation theory is employed in DR asymmetrical structure to improve the impedance and AR bandwidth of the antenna. Two DRs are taken, and calculate the volume (Vdra) of DR1 and DR2 are 140 and 384 (total = 524) mm3, as shown in figure 1c. The total volume (Vtotal) of DRA is 728 mm3 (13 × 7 × 8). So air-spaced volume Vs = 728–524 = 204 mm3. The effective permittivity (εreff) of the DR depends on the volume (Vs, Vdra) and εDR = 10.2. With and without perturbation, DR is shown in figure 2. So calculated εreff = 7.14 from (1) and quality factor Q and Qp are 2.1 and 1.79 using Eqs. (2)–(3). Thus, as the εreff and Qp of DR decreases vice versa, bandwidth of antenna is improved.

$$ \varepsilon_{reff} = \frac{{\varepsilon_{DR} Vdra + \varepsilon_{0} Vs}}{Vdra + Vs} $$
(1)
$$ {\text{Q = }}\frac{{{1 + 3}\left( {\frac{\pi }{{\sqrt {\varepsilon_{DR} } }}} \right)^{2} }}{{\left( {\frac{\pi }{{\sqrt {\varepsilon_{DR} } }}} \right)^{3} \left[ {{1 + }\left( {\frac{\pi }{{\sqrt {\varepsilon_{DR} } }}} \right)^{2} } \right]}} $$
(2)
$$ {\text{Qp = }}\frac{{{1 + 3}\left( {\frac{\pi }{{\sqrt {\varepsilon_{reff} } }}} \right)^{2} }}{{\left( {\frac{\pi }{{\sqrt {\varepsilon_{reff} } }}} \right)^{3} \left[ {{1 + }\left( {\frac{\pi }{{\sqrt {\varepsilon_{reff} } }}} \right)^{2} } \right]}} $$
(3)

The total size of DRA is 13 mm × 7 mm × 8 mm. Firstly, the DRA key parameters are calculated using eq. (4), kx, ky, kz, k0, and resonant frequency values are 448.79, 392.69, 196.34, 230.79 and 10.65 GHz. Similarly, DR1 & DR2 resonant frequencies are 12.7 & 10.02 GHz. The simulated resonant frequency of DRA is 11.28 GHz.

$$ {\text{f}}_{{{\text{TE}}_{{{\text{lnm}}}} }} = \frac{\nu }{{2\pi \sqrt {\varepsilon_{{{\text{reff}}}} } }}\sqrt {{\text{k}}_{{\text{x}}}^{2} + {\text{k}}_{{\text{y}}}^{2} + {\text{k}}_{{\text{z}}}^{2} } $$
(4)
$$ k_{x} = \frac{l\pi }{{l_{{{\text{eff}}}} }}, {\text{k}}_{y} = \frac{{{\text{n}}\pi }}{{w_{{{\text{eff}}}} }}, {\text{k}}_{z} \tan \left( {k_{z} \frac{{h_{{{\text{DR}}}} }}{2}} \right) = \sqrt {\left( {\varepsilon_{{{\text{reff}}}} - 1} \right)k_{0}^{2} - {\text{k}}_{z}^{2} } $$
(5)
$$ l_{{{\text{eff}}}} { = }l_{{{\text{dr1}}}} {\text{ + l}}_{{{\text{dr2}}}} \; {\text{and}}\;w_{{{\text{eff}}}} {\text{ = w}}_{{{\text{dr1}}}} {\text{ + w}}_{{{\text{dr2}}}} $$
(6)

where k0 represents wave number in the free space, kx, ky, and kz are wavenumbers.

2.3 Field modes and CP mechanism

The simulated E-field vector configuration has been analyzed by field rotation inside the proposed asymmetric DR at the different frequencies. The concept of E-field analysis is reported in [3, 6, 12, 17]. Eq. (1), (4)–(6) are used to calculate TE111 fundamental as well as the dominant mode of the antenna. Figure 5 shows the impedance matching of the antenna with the resonant frequencies of the different operating modes. There are two peaks in the real part of the impedance plot in the antenna's passband, confirming the antenna's operation with two resonating modes at frequencies 9.1 and 11.28 GHz. Figure 6 shows the E-field distribution inside the DR at 9.1 and 11.28 GHz [17, 19,20,21,22]. Figures 6(a)–(b) confirm the operation of an antenna with the fundamental and third-order mode, i.e., \(TE_{\delta 11}\) and \(TE_{\delta 13}\) at frequency 9.1 and 11.28 GHz, respectively. For understanding the mechanism of CP generation, the E-field distribution is also analyzed in the constant \(z = h\) plane at different time instants, shown in figures 7(a)–(d). The rotation of the field vector in the clockwise direction represents the dominant left-hand CP field radiation that could be verified with the radiation pattern analysis (figure 15) reported in the later section of this manuscript. The E-field distribution in the constant z-plane is stated at the first-order mode frequency. However, the analysis also represents the clockwise rotation of the field vectors at the higher-order mode frequency. The E-field distribution and the radiation mechanism of the proposed CPDRA are reported broad AR bandwidth [22,23,24].

Figure 5
figure 5

The impedance plot of the antenna.

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Figure 6
figure 6

The E-field distribution at (a) 9.1 and (b) 11.28 GHz.

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Figure 7
figure 7

The E-field distribution in constant \(z = h\) plane at frequency 9.1 GHz at the time (a) \(t = 0,\) (b) \(t = T/4\), (c) \(t = T/2\) and (d) \(t = 3T/4\).

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3 Parametric study

The parametric study has been performed to obtain optimum dimensions for achieving the best results of the DRA and for which parameter to give the CP inside the antenna structure.

3.1 Variation of the height of DR

The effect of DR heights as tdr1 and tdr2 are varied, and it observed from figures 8 (a)–(d) that the best values are tdr1 = 4 mm and tdr2 = 8 mm for widen impedance bandwidth and AR bandwidth, respectively.

Figure 8
figure 8

Reflection coefficient and Axial Ratio for various height (tDR1 (a and b) and tDR2 (c and d)) of DR.

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3.2 Variation parameter in F-shaped strip

The F shaped strip with main key parameters are st1, st2, lt1, lt2, and lt3. The best values are chosen through optimization as in figures 9 (a–b) on st1 = 6 mm (− 3 to 3) and figure 9 (c–d) on st2 = 2 mm (− 10 to − 8).

Figure 9
figure 9

Reflection coefficient and Axial Ratio for varies st1 (a–b), and st2 (c–d).

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3.3 Variation of the partial ground (Lg)

The proposed DRA has with or without an F-shaped metallic strip in the partial ground to achieve the UWB range. Variation in the Lg parameter is shown in figure 10, and the best value of Lg = 9.4 mm (− 12.5 to − 3.1) is chosen.

Figure 10
figure 10

Reflection coefficient with varies Lg.

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The proposed compact DRA is compared with other literature-reported structures related to parameters such as antenna size, peak gain, impedance bandwidth and AR bandwidth for ultra-wideband operation, as shown in Table 3.

Table 3 Comparison of CP DRA.
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4 Results and discussions

The fabricated prototype with testing and measurement setup is shown in figures 11 (a–b), and the fabricated DRA view is shown in figures 11 (c–d). The simulated and measured S11 plot and AR plot of the designed antenna are presented in figures 12 (a–b). The measured S11 is almost similar to the simulated ones. A slight left-shifting of the three resonance frequencies is observed due to the favitite epoxy adhesive, which is used to join the DR layers on the substrate. Simulated and measured impedance bandwidths of 63.8% (6.4 to 12.4 GHz) and 66.7% (5.9 to 11.8 GHz) have been reported at a central frequency of 9.35 GHz and 8.85 GHz, respectively. Subsequently, from figure 12b, the simulated AR bandwidth is obtained 63.8% (6.47 to 12.3 GHz) and measured are 2.52% and 43.7% (5.8 to 6.1 GHz and 6.9 to 10.9 GHz).

Figure 11
figure 11

(a–b) Measurement setup back and Front view of DRA (c–d) Back and front view of fabricated DRA.

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Figure 12
figure 12

Simulated and measured; (a) S11 and (b) AR w.r.t frequency.

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Figure 13(a) shows the simulated gain at 11 GHz. The simulated and measured gain are shown in figure 13b. The antenna's simulated peak gain is 6.01dBi at 11.5 GHz, while the measured peak gain is 5.6 dBi at 11 GHz, according to the figure 13(b).

Figure 13
figure 13

(a) 3-D simulated gain plot at 11 GHz and (b) Gain w.r.t. frequency.

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This proposed antenna is dominated by the left-handed polarization compared to the right-handed polarization, as shown in figure 14. Figures 15(a–b) showed the measured and simulated radiation patterns of the proposed antenna at 6.6 and 9.1 GHz for XZ and YZ planes. A good agreement reported between simulated and measured results. Referring to figures 15(a–b), it is observed that the LHCP fields are much stronger in comparison to the RHCP for CP DRA.

Figure 14
figure 14

Polarization (dB) of the proposed antenna as LHCP and RHCP with respect to frequency.

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Figure 15
figure 15

Radiation pattern (a) LHCP and RHCP at 6.6 GHz, (b) LHCP and RHCP at 9.1 GHz.

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5 Conclusion

A compact asymmetric structured DR is designed and fabricated for CP UWB applications. The antenna is used for two DR blocks with asymmetric height using QWT feeding and an F-shaped metallic strip in the ground plane to obtain simulated AR bandwidth of 63.8% and measured AR bandwidth of 43.7% in the frequency range from 6.4 to 12.4 GHz for circularly polarized DRA. These attractive performances make the present design suitable for UWB operations. This antenna can be used for (C and X band) multiple applications like military applications, satellite communication, weather forecasting, and defense tracking system.