Design, modeling and analysis of dual-feed defected ground microstrip patch antenna with wide axial ratio bandwidth

Abstract

In this paper, a low profile microstrip patch antenna with defected ground structure is proposed for wide impedance and axial ratio bandwidth. The antenna structure consists of two feed lines with a circular defect loaded in the ground surface. The proposed antenna shows an impedance bandwidth of 71.92% ranging from 8.95 to 19 GHz covering X, Ku and K band applications. The mutual coupling between the two feed lines is suppressed by means of a single and double arc-shaped defect embedded in the ground plane. The mutual coupling is suppressed to a level of − 37.5 dB with improved radiation performance. The measured 3-dB axial ratio bandwidth of the proposed antenna ranges from 8.85 to 11 GHz. An equivalent circuit model of the designed antenna is also proposed for theoretical analysis and theoretical results are verified with simulated and measured results.

1 Introduction

Due to the rapid expansion of wireless communication, there is an increasing demand for antennas that provide high gain as well as polarization diversity along with wideband operation. To fulfill all these requirements, considerable attention has been given to circularly polarized (CP) antennas because they have the ability to suppress the detrimental effect of multipath fading and also they allow the signal reception caused by misalignment of receiving and transmitting antennas. CP planar antennas are most popularly used nowadays in portable handheld terminals because of their low profile, small size, cheap cost and easy integrability with handheld devices. There is no problem of orientation between transmitting and receiving elements involving CP antennas. CP antennas resolve the problem of fading leading to a system with better spectral efficiency and more throughput. A classical method for stimulating circular polarization involves excitation of two linearly polarized orthogonal modes of identical amplitude with quadrature phase difference between them.

A CP band can be excited by single feed or dual feed. A number of techniques have been presented by different researchers for the excitation of circular polarization by means of cutting a portion of the square radiator, inserting a slot, slit or stub in the ground or patch, by loading an active device in the patch, etc. [1,2,3,4,5,6,7,8,9,10]. A patch antenna with unobtrusive single feed and loaded metamaterial is proposed in Ref. [1]. This antenna consists of right/left-handed mushroom-shaped resonators for exciting circular polarization. In Ref. [2], circular polarization is achieved by loading parasitic shorting element. In Ref. [3], a single feed patch antenna loaded with U-slot and truncated corner square patch is proposed for circular polarization radiation. Single feed techniques provides a narrow axial ratio bandwidth. To achieve a broad bandwidth, a dual-fed L-shaped structure with cavity backing is reported in Ref. [4]. The ground plane with cavity backing helps in improving the coupling factor thus increasing the antenna bandwidth. CP can also be achieved by parasitic loading [5] or by inserting stubs [6]. Dual band CP operation is realized by introducing a parasitic patch under the radiating patch in Ref. [5] whereas in Ref. [6] L-shaped stubs are loaded exterior to the trimmed patch so that outer and inner mode can be excited. By introducing asymmetrical slits on each corner of square patch, circular polarization is achieved in Ref. [7, 8]. For size reduction and to acquire circular polarization, arrowhead-shaped slotted antenna is proposed in Ref. [9]. The Koch fractal geometry can also be used for circular polarization [10]. The dual-feed configuration provides wider axial ratio bandwidth keeping quadrature phase difference between the two feed elements. However, the major problem lying with dual-feed antennas is mutual coupling between the two feed elements.

The problem of mutual coupling can be solved by using defected ground structure technique consequently improving the antenna performance parameters. In the last decade, a number of studies have been reported for reducing the mutual coupling effect using ground with induced defects. In Ref. [11], a circular ring is embedded in the ground surface for reducing the mutual coupling effect. In Ref. [12] the ground plane is loaded with a dumbbell shaped slot. A spiral-shaped ground with a defect is proposed in Ref. [13]. Finally, an inverted defective U-shaped ground in discussed in Ref. [14]. In this paper, we propose a dual-feed microstrip antenna with wide impedance and axial ratio bandwidth for X, Ku and K band satellite applications. The two microstrip line-feeds excites two orthogonal waves of same amplitude with a 90-degree phase difference between them for circular polarization. The mutual coupling effect between the feeds is minimized by integrating a pair of circular arc in the ground surface of the designed antenna. The proposed antenna shows a − 10 dB impedance bandwidth of 71.92% ranging from 8.95 to 19 GHz and 3-dB axial ratio bandwidth from 8.85 to 11 GHz. The main advantage of proposed antenna is its low profile and simple geometry without any loading of active devices, truncation at corners or perturbations in the patch for exciting CP radiations. Moreover, the proposed antenna structure has been theoretically analyzed using the cavity model and equivalent circuit approach.

2 Antenna design

The antenna consists of a microstrip line of length $L_\mathrm{ml}$ and width $w_\mathrm{ml}$ with circular slot of radius a embedded in the ground plane underneath the open end of microstrip line as shown in Fig. 1. The antenna is referred as Ant1 and the total size of antenna is $D\times D \times h~\hbox {mm}^{3}$. In the second step, one more microstrip line of same dimension (Ant1) is placed orthogonal to the first feed line and designed antenna is referred as Ant2. The dual-fed circular slot defected ground antenna schematic and fabricated prototype is shown in Fig. 2. The two feeds are represented as feed1 and feed2. The overall size of designed antenna is $B\times B \times h~\hbox {mm}^{3}$.

Fig. 1

Design layout of Ant1 a schematic, b top view of fabricated prototype, c bottom view of fabricated prototype

Fig. 2

Design layout of Ant2 a schematic, b top view of fabricated prototype, c bottom view of fabricated prototype

Further, in order to achieve circular polarization in the designed antenna (Ant2), the length of feed2 is increased by a deference of quarter wavelength, thus introducing a phase shift of 90 degree between the resonating modes. The designed structure with asymmetric length dual-feed arrangement is referred as Ant3. Figure 3a, b shows the schematic and fabricated prototype of proposed Ant3 with overall antenna size $B \times B_{1} \times h~\hbox {mm}^{3}$ respectively. Additionally, to reduce the effect of mutual coupling between the feed lines, a circular arc-shaped slot of width s is embedded in the ground plane at a distance of d from the center of circular defect. The dual-fed circular arc-shaped slot loaded defected ground antenna schematic and fabricated prototype is shown in Fig. 3c, d, respectively, and is referred as Ant4. To further reduce the mutual coupling effect between the feed lines, one more slot (which is a mirror image of circular arc-shaped slot embedded in Ant4) is loaded in the designed antenna. The dual-feed antenna loaded with two arc-shaped slots and circular defect in ground surface is referred as Ant5. Figure 3e, f shows the schematic and prototype of proposed Ant5, respectively. The simulation and optimization of proposed antenna prototype are carried out using finite element method based commercially available software package, Ansys HFSS. The detailed dimensions of the proposed structure are listed in Table 1.

Fig. 3

Design layout of the proposed antennas a schematic of Ant3, b fabricated prototype of Ant3 (top view), c schematic of Ant4, d fabricated prototype of Ant4 (bottom view), e schematic of Ant5, f fabricated prototype of Ant5 (bottom view)

3 Theoretical considerations

The equivalent circuit of microstrip line-fed defected ground antenna is illustrated in Fig. 4. Microstrip line of 50 $\Omega$ is used for feeding the antenna structure. The ground plane embedded circular defect radius a can be calculated as [15]

$$\begin{aligned} a=\frac{F}{\left\{ {1+\frac{50h}{\pi \varepsilon _\mathrm{r} F}\left[ {\ln \left( {\frac{\pi F}{50h}} \right) +1.7726} \right] } \right\} ^{1/2}} \end{aligned}$$

(1)

where h and $\varepsilon _\mathrm{r}$ are the thickness and dielectric constant of the substrate, respectively. The value of F is calculated as

$$\begin{aligned} F=\frac{8.791\times 10^{9}}{f_\mathrm{min} \sqrt{\varepsilon _r }} \end{aligned}$$

(2)

where $f_\mathrm{min}$ is the design frequency of the wideband microstrip antenna. A deference of quarter wavelength is present in feed lines length for the excitation of circular polarization. The length $L_\mathrm{ml}$ of feed1 and $L_\mathrm{ml2}$ of feed2 are considered in the multiple of quarter wavelength and are calculated as

$$\begin{aligned}&L_\mathrm{ml} =l_1 +l_2 =\frac{5}{4}{\lambda }_\mathrm{eff}\end{aligned}$$

(3)

$$\begin{aligned}&L_\mathrm{ml2} =\frac{5}{4}{\lambda }_\mathrm{eff} +\frac{1}{4}{\lambda }_\mathrm{eff} =\frac{3}{2}{\lambda }_\mathrm{eff} \end{aligned}$$

(4)

where

$$\begin{aligned} \lambda _\mathrm{eff} =\frac{c}{f_\mathrm{min} \sqrt{\varepsilon _\mathrm{eff} }} \end{aligned}$$

(5)

where c is the speed of light in free space and $\varepsilon _\mathrm{eff}$ is the effective dielectric constant of the microstrip line [15]

$$\begin{aligned} \varepsilon _\mathrm{eff} =\frac{\varepsilon _\mathrm{r} +1}{2}+\frac{\varepsilon _\mathrm{r} -1}{2}\frac{1}{\sqrt{1+12h/w_\mathrm{ml} }} \end{aligned}$$

(6)

where $w_\mathrm{ml}$ is the width of microstrip line.

To calculate the position of circular defect embedded in the ground plane, the microstrip line has been divided into two parts: length $l_{1}$ and length $l_{2}$. A lossy microstrip line can be modeled as a tank circuit with parameters series resistance R, series inductance L,  shunt conductance G and shunt capacitance C. These parameters R, L, G and C are in per unit length and can be calculated for length $l_{1}$ and $l_{2}$ for microstrip line. The parameters $R_{11}$, $L_{11}$, $G_{11}$, $C_{11}$ and $R_{12}$, $L_{12}$, $G_{12}$, $C_{12}$ correspond to the microstrip line of length $l_{1}$ and $l_{2}$ of feed1, respectively. Similarly, the parameters $R_{21}$, $L_{21}$, $G_{21}$, $C_{21}$ and $R_{22}$, $L_{22}$, $G_{22}$, $C_{22}$ correspond to microstrip line of length $l_{1}$ and $l_{2}$ of feed2, respectively. The microstrip line impedance $Z_{11}$ and $Z_{12}$ of feed1 and $Z_{21}$ and $Z_{22}$ of feed2 offered by length $l_{1}$ and $l_{2}$, respectively, are given as

$$\begin{aligned} Z_{11}= & {} \sqrt{\frac{R_{11} +j\omega L_{11} }{G_{11} +j\omega C_{11}}} \end{aligned}$$

(7)

$$\begin{aligned} Z_{12}= & {} \sqrt{\frac{R_{12} +j\omega L_{12} }{G_{12} +j\omega C_{12} }} \end{aligned}$$

(8)

$$\begin{aligned} Z_{21}= & {} \sqrt{\frac{R_{21} +j\omega L_{21} }{G_{21} +j\omega C_{21} }} \end{aligned}$$

(9)

$$\begin{aligned} Z_{22}= & {} \sqrt{\frac{R_{22} +j\omega L_{22} }{G_{22} +j\omega C_{22} }} \end{aligned}$$

(10)

Table 1 Design specifications for proposed antennas

A fringing capacitance $C_\mathrm{f}$ gets introduced at the open-ended discontinuity of the microstrip line which is calculated as [16]

$$\begin{aligned} C_f =\frac{\Delta l\sqrt{\varepsilon _\mathrm{eff} }}{cZ_0 } \end{aligned}$$

(11)

where $\Delta l$ is the extended fringing field length at the end of microstrip line and $Z_{0}$ is the characteristic impedance of microstrip line. The impedance offered by fringing capacitance of feed1 and feed2 referred as $Z_\mathrm{f1}$ and $Z_\mathrm{f2}$ respectively which is determined as

$$\begin{aligned} Z_\mathrm{f1} =Z_\mathrm{f2} =\frac{1}{j\omega C_\mathrm{f} } \end{aligned}$$

(12)

The coupling between the two circuits can be modeled by the help of coupling capacitance $C_\mathrm{p}$ given by [17]

$$\begin{aligned} C_\mathrm{p} =\frac{-\left( {C_1 +C_2 } \right) +\sqrt{\left( {C_1 +C_2 } \right) ^{2}-4C_1 C_2 \left( {1-C_k^{-2} } \right) }}{2} \end{aligned}$$

(13)

Fig. 4

Modeling of DGS with microstrip line

where $C_{1}$ and $C_{2}$ are the capacitances offered by both the circuits. The coupling coefficient $C_\mathrm{k}$ between the two networks is

$$\begin{aligned} C_\mathrm{k} =\frac{1}{\sqrt{Q_1 Q_2 }} \end{aligned}$$

(14)

where $Q_{1}$ and $Q_{2}$ are the quality factor in both the networks. The coupling capacitance between feed1 and circular defect is referred as $C_\mathrm{p1}$. Similarly, the coupling capacitance between feed2 and circular defect is referred as $C_\mathrm{p2}$. To model the coupling between fringing fields due to two feed lines, a coupling capacitance $C_\mathrm{ff}$ is introduced in the circuit model. A coupling capacitance $C_\mathrm{PC}$ between both the feed lines is also introduced. The coupling capacitance values may be calculated by the help of Eq. (13). The quality factor of microstrip line is determined as

$$\begin{aligned} Q=\omega \frac{L}{R} \end{aligned}$$

(15)

The coupling capacitance $C_\mathrm{p1}$, $C_\mathrm{p2}$, $C_\mathrm{ff}$ and $C_\mathrm{PC}$ offers $Z_\mathrm{p1}$, $Z_\mathrm{p2}$, $Z_\mathrm{f}$ and $Z_\mathrm{PC}$ impedance, respectively, which can be calculated as

$$\begin{aligned}&Z_\mathrm{P1} =\frac{1}{j\omega C_\mathrm{p1} };\;Z_\mathrm{P2} =\frac{1}{j\omega C_\mathrm{p2} };\,Z_\mathrm{f} =\frac{1}{j\omega C_\mathrm{ff} };\;\nonumber \\&Z_\mathrm{PC} =\frac{1}{j\omega C_\mathrm{PC} } \end{aligned}$$

(16)

The coupling inductance $L_\mathrm{PL}$ between the two microstrip lines is determined from [17]

$$\begin{aligned} L_\mathrm{PL} =\frac{{C_\mathrm{k}} ^{2}\left( {L_1 +L_2 } \right) +\sqrt{{C_\mathrm{k}} ^{2}\left( {L_1 +L_2 } \right) ^{2}+4{C_\mathrm{k}} ^{2}\left( {1-{C_\mathrm{k}} ^{2}} \right) L_1 L_2 }}{2\left( {1-{C_\mathrm{k}} ^{2}} \right) } \end{aligned}$$

(17)

where $L_{1}$ and $L_{2}$ represents the inductance of feed1 and feed2, respectively. The impedance offered by coupling inductance $L_\mathrm{PL}$ is calculated as

$$\begin{aligned} Z_\mathrm{PL} =j\omega L_\mathrm{PL} \end{aligned}$$

(18)

The input impedance $Z_\mathrm{slot}$ of embedded ground slot is modeled by an ideal transformer with microstrip line as shown in Fig. 4. The turn ratio N of transformer is determined as [18]

$$\begin{aligned} N=\sqrt{\frac{Z_0 }{Z_\mathrm{slot} }} \end{aligned}$$

(19)

The slot impedance $Z_\mathrm{slot}$ of circular defect is calculated from [19]

$$\begin{aligned} Z_\mathrm{slot} =\frac{\eta ^{2}}{4Z_\mathrm{cy}} \end{aligned}$$

(20)

where $\eta$ is the intrinsic impedance, $Z_\mathrm{cy}$ is input impedance of the cylindrical dipole given as

$$\begin{aligned} Z_\mathrm{cy} =R_\mathrm{r} \left( { kl} \right) -j\left[ {120\left( {ln\frac{l}{w}-1} \right) \cot \frac{kl}{2}} \right] \end{aligned}$$

(21)

where $R_\mathrm{r}$ is defined as the radiation resistance of the dipole determined as

$$\begin{aligned} R_\mathrm{r} =\frac{\eta _0 }{2\pi } \int _0^\pi \left[ {\frac{\left[ {\cos \left( {\frac{k^{2}}{2}\cos \theta } \right) -\cos \left( {\frac{kl}{2}} \right) } \right] ^{2}}{\sin \theta }} \right] \mathrm{d}\theta \end{aligned}$$

(22)

The slot is integrated in the ground plane; thus, the input impedance of the slot is added to the circuit model by an ideal transformer with turn ratio N determined by

$$\begin{aligned} Z{\prime }_\mathrm{slot} =NZ_\mathrm{slot} \end{aligned}$$

(23)

Figure 5 shows the equivalent circuit model for the proposed antenna structure. The total input impedance $Z_\mathrm{in}$ of the model is given by

$$\begin{aligned} Z_\mathrm{in}= & {} \left[ \left( {\frac{1}{\left( {Z_{11} +Z_{12} } \right) }+\frac{1}{Z_\mathrm{P1} }++\frac{1}{Z_\mathrm{f1} }+\frac{1}{Z_\mathrm{slot}^{\prime } }} \right) \right. \nonumber \\&+\,\left. \left( {\frac{1}{Z_\mathrm{PL} }+\frac{1}{Z_\mathrm{f} }+\frac{1}{Z_\mathrm{PC} }} \right) \right. \nonumber \\&+\,\left. \left( {\frac{1}{\left( {Z_{21} +Z_{22} } \right) }+\frac{1}{Z_\mathrm{P2} }+\frac{1}{Z_\mathrm{f2} }} \right) \right] ^{-1} \end{aligned}$$

(24)

The $S_{11}$ value of proposed antenna is calculated as

$$\begin{aligned} S_{11}=20\log \left( {\left| \hbox {r} \right| } \right) \end{aligned}$$

(25)

Fig. 5

Equivalent circuit model for the proposed antenna structure

where r is the reflection coefficient and is

$$\begin{aligned} \hbox {r}=\frac{Z_\mathrm{in} -Z_0 }{Z_\mathrm{in} +Z_0 } \end{aligned}$$

(26)

The voltage standing wave ratio (VSWR) can be calculated as

$$\begin{aligned} \hbox {VSWR}=\frac{1+\left| \hbox {r} \right| }{1-\left| \hbox {r} \right| } \end{aligned}$$

(27)

The parameters for each port can be calculated using Eqs. (24–27) i.e. $Z_\mathrm{in1}$, $S_{11}$, $\hbox {r}_{1}$, $\hbox {VSWR}_{1}$ and $Z_\mathrm{in2}$, $S_{22}$, $\hbox {r}_{2}$, VSWR$_{2}$ corresponding to port 1 and port 2, respectively. The parameter $S_{12}$ or $S_{21}$ is related with other parameters by following equation [20]

$$\begin{aligned} Z_\mathrm{in1}= & {} Z_0 \frac{\left( {1+S_{11} } \right) \left( {1-S_{22} } \right) +S_{21} ^{2}+2S_{21} \frac{\hbox {VSWR}_2 \left( {1-S_{22} } \right) -\hbox {VSWR}_1 S_{21} }{\hbox {VSWR}_1 \left( {1-S_{11} } \right) -\hbox {VSWR}_2 S_{21} }}{\left( {1-S_{11} } \right) \left( {1-S_{22} } \right) -S_{21} ^{2}}\nonumber \\ \end{aligned}$$

(28)

The parameter $S_{21}$ can be easily calculated by Eq. (28) after calculating $Z_\mathrm{in}$, $S_{11}$/$S_{22}$ and VSWR using Eqs. (24), (25) and (27), respectively.

4 Results and discussion

The substrate RT Duriod 5880 is used for the fabrication of all proposed antenna designs. The dielectric constant $\varepsilon _\mathrm{r}$, loss tangent $tan~\delta$ and height h of the substrate are 2.2, 0.0009 and 0.762 mm, respectively. The S-parameters of the fabricated antennas are measured using Agilent Vector Network Analyzer PNA L series. The theoretical, simulated and measured return loss behavior of Ant1 is shown in Fig. 6 and is considered as a reference antenna for the present study. Ant1 shows two resonances at 10.8 and 16 GHz with wide impedance bandwidth ranging from 9.8 to 17.55 GHz.

Fig. 6

Theoretical, simulated and measured S-parameters of Ant1

Fig. 7

Theoretical, simulated and measured S-parameters of Ant2

The Ant2 with microstrip line dual-feed mechanism shows an impedance bandwidth ranging from 9.1 to 14.95 GHz with single resonance at 10.15 GHz. The S-parameter variation of Ant2 is shown in Fig. 7, and an impedance bandwidth of 48.65% is achieved in this antenna. The mutual coupling between the two feed lines degrades the antenna performance resulting in a smaller − 10 dB impedance bandwidth and single resonance in Ant2 compared to Ant1. Because of a small distance between the open ends of two microstrip lines, a large amount of mutual coupling will be present among the feed lines. The $S_{21}$ of Ant2 is about − 12.87 dB at 12 GHz. The theoretical, simulated and measured S-parameters of Ant2 are in good agreement with each other. However, compared to simulated and measured results, the theoretical $S_{11}$ variation shows better resonance at 11.4 GHz. This can be understood by the concept of current distribution within the metal strip and the ground plane. The presence of ground plane under the feed line results in an unequal division of current in the metal conductor and the ground surface. Because of this the normalized series resistance offered by the strip and the ground are of different values. Since there is no ground plane under the length $l_{2}$, the normalized series resistance offered by this segment is negligible.

The theoretical, simulated and measured S-parameters of Ant3 are shown in Fig. 8. There is a difference of $\lambda /4$ between the two rectangular feed lines in Ant3. Ant3 shows an impedance bandwidth of 52.53% ranging from 9.1 to 15.25 GHz. The mutual coupling $S_{21}$ between the two feed lines is almost same as in Ant2. In Fig. 8, the measured and simulated $S_{11}$ variations are in good match with theoretical results.

Fig. 8

Theoretical, simulated and measured S-parameters of Ant3

Fig. 9

Simulated and measured $S_{11}$ variation with frequency of Ant4 ($d=6\, \mathrm{mm}$ and $s=1\, \mathrm{mm})$)

Fig. 10

$S_{11}$ variation with frequency of Ant4 for different arc width

Fig. 11

$S_{11}$ variation with frequency of Ant4 for different arc position

Fig. 12

$S_{21}$ variation with frequency of Ant4 for different arc width

Fig. 13

$S_{21}$ variation with frequency of Ant4 for different arc position

Fig. 14

Simulated and measured S-parameters of Ant5

Fig. 15

Simulated and measured axial ratio of Ant3, Ant4, Ant5

Fig. 16

Simulated and measured gain of Ant2, Ant3, Ant4, Ant5

Figure 9 shows simulated and measured return loss comparison for Ant4. The − 10 dB impedance bandwidth ranges from 8.95 to 19 GHz (71.92%). The parametric comparisons of $S_{11}$ with frequency for different values of circular arc width s and radius d are shown in Figs. 10 and 11, respectively. The mutual coupling comparisons for different values of circular arc width s and radius d are shown in Figs. 12 and 13, respectively. In the figure, it can be observed that the value of $S_{21}$ is suppressed to a level of − 20.3 dB for $s = 1\, \hbox {mm}$ and $d = 6\, \hbox {mm}$.

The S-parameters variation with frequency for Ant5 is shown in Fig. 14. Ant5 shows an impedance bandwidth of 71.92% ranging from 8.95 to 19 GHz. By embedding two circular arcs in the ground plane, the mutual coupling value can be suppressed up to a level of − 37.5 dB in the band of operation. The simulated results of proposed antenna are in good match with measured results. The $S_{22}$ and $S_{12}$ variations of all the designed antennas are similar to their relative $S_{11}$ and $S_{21}$ variations; therefore, in order to avoid the repetition of figures, $S_{11}$ and $S_{21}$ parameters have been shown only.

Fig. 17

Simulated and measured Co-Polar (CP) and Cross-Polar (XP) radiation pattern at 10 GHz a Ant2 in XY plane, b Ant2 in XZ plane, c Ant3 in XY plane, d Ant3 in XZ plane, e Ant4 in XY plane, f Ant4 in XZ plane, g Ant5 in XY plane, h Ant5 in XZ plane

Figure 15 illustrates the axial ratio variation with frequency of Ant3, Ant4 and Ant5. For creating a 90-degree phase difference between the resonating modes, a pair of feed lines with a difference of quarter wavelength between the two are used. Ant3 shows circular polarization ranging from 9.1 to 10 GHz. The 3-dB axial ratio value of Ant4 ranges from 9.15 to 10.35 GHz with a wide band of more than 1.2 GHz. The measured 3-dB axial ratio bandwidth of Ant5 ranges from 8.85 to 11 GHz with a wide bandwidth of more than 2.1 GHz. Thus, Ant5 provides largest 3-dB axial ratio bandwidth among the proposed dual-feed antenna designs.

The gain of proposed antennas Ant2, Ant3, Ant4 and Ant5 is shown in Fig. 16. Ant2 and Ant3 show peak gain of 5.2 and 5.4 dBi at resonance frequencies 9.5 and 16.6 GHz, respectively. The average gain of Ant2 and Ant3 is about 5.2 and 5 dBi, respectively, within the radiating band. An increase in the value of gain is observed in Ant4 at the frequency 16 GHz due to the isolation effect provided by the embedded circular arc. A further improvement in antenna gain is observed in Ant5 by means of a double arc-shaped embedded defect in the ground surface. The peak values of measured gain in Ant4 and Ant5 are 5.9 and 6.9 dBi, respectively. A small difference in simulated and measured results is present due to conventional photolithography fabrication process, soldering of SMA connectors and cable effects. Figure 17 shows the Co-Polar (CP) and Cross-Polar (XP) radiation characteristics of presented antennas in XY and XZ planes. For comparison purpose, all radiation patterns are plotted at 10 GHz. Ant2 and Ant3 show higher XP level because of dual feed, whereas XP level is suppressed in Ant4 and Ant5 because of the use of DGS. Ant5 shows best radiation characteristics with minimum XP level and better isolation between CP and XP radiations.

5 Conclusion

A dual-feed microstrip patch antenna with wide − 10 dB impedance and 3-dB axial ratio bandwidth is presented. The dual feeds stimulate the two orthogonal waves with 90-degree phase difference among them to obtain circular polarization without cutting or notching any region of the radiating patch. An equivalent circuit model is also proposed for the antenna structure, and it is found that theoretical and simulated results are a good match with measured results. Further, single and double arc-shaped DGS are embedded in the ground plane to reduce the mutual coupling effect between the feed lines. The antenna shows a wide impedance bandwidth of 71.92% ranging from 8.95 to 19 GHz with a measured peak gain of around 6.9 dBi in the resonating band. The proposed antenna does not involve truncation of corners or any loading of active devices used for the excitation of circular polarization. Compared with the other existing wideband CP antennas, the presented antenna is very easy to fabricate as well as has wide impedance and axial ratio bandwidth, thus making it a potential candidate for the current wireless applications.