Design of RFID Tag Antenna with Impedance Matching Techniques at UHF Band

Abstract

A suitable antenna is going to be designed for RFID (Radio Frequency Identification) tag system. Long read range is possible if antenna impedance is well matched to chip impedance. This paper deals with the design and analysis techniques of a basic antenna at the required narrowband of frequency (865.6–867.6 MHz) and extended to impedance matching by different techniques such as parasitic, T-stub, shorting pin, etc. The proposed techniques are designed in Ansoft HFSS13.0 and measured with network analyzer which is suitable for few commercial chips and applications.

1 Introduction

The microstrip antenna plays a vital role in communication world for four decades. Microstrip antenna is suitable for different applications like RFID tag, wireless communication, space communication, and biomedical, etc., due to their less weight, low-cost, and narrow bandwidth.

RFID tag is a device having transceiver, used for identification of objects which are made with metal based from few feet to hundreds of feet in direct contact. RFID tags are working in narrowband characteristics. These are classified into active RFIDs, passive RFIDs, and semi-active RFIDs [1,2,3]. RFID tags are working at different frequency ranges from low frequency to UHF. An antenna plays important role in RFID tag to transmit the signal and receive the reflected signal from the object.

The antenna to be designed is in the range 865–880 MHz which covers different spectral bands [4] used in several countries like India (865–867 MHz), Iran (865–868 MHz), South Africa (865.6–867.6 MHz), Singapore (866–869 MHz), etc. The basic RFID working principle is shown in Fig. 1. Designing of different antennas, enhancement techniques of inductive impedance of an antenna, and read range calculations with different commercially available chips are discussed in the following sections.

Fig. 1

Basic RFID design

2 RFID Tag Antenna Design Approach

To design a suitable RFID tag antenna, the important parameter to be considered is

• (i) Read Range Equation

The most important characteristic of a passive RFID tag is the read range. The maximum distance of RFID reader is able to detect the modulated backscattered signal and successfully identify a tag. The read range may be roughly calculated using the FRIIS transmission formula [5,6,7,8]

$$R_{\hbox{max} } = \frac{\lambda }{4\pi }\sqrt {\frac{{P_{\text{t}} G_{\text{t}} G_{\text{r}} \left( {1 - \left|\Gamma \right|^{2} } \right)\rho }}{{P_{\text{th}} }}}$$

(1)

where

$\lambda$ :

length

$P_{\text{t}}$ :

Effective radiated power (i.e., 0.5, 2, and 4 W)

$G_{\text{t}}$ :

Gain of reader antenna (transmitter)

$G_{\text{r}}$ :

Gain of tag antenna (receiver)

$P_{\text{th}}$ :

Minimum threshold power which can activate the circuit device (i.e., −10, −15, −18, and −20 dBm)

$\rho$ :

Polarization mismatch between the reader and tag antenna

$\Gamma$ :

Voltage reflection coefficient

$\left|\Gamma \right|^{2}$ :

Power reflection coefficient

$\left( {1 - \left|\Gamma \right|^{2} } \right)$ :

Transmission coefficient

• (ii) Impedance matching condition

According to the maximum power transfer theorem, tag antenna impedance is a complex conjugate of a chip impedance [8] and the circuit is given in Fig. 2. The voltage reflection coefficient is calculated from Eq. 2 [7, 9],

$$\varGamma = \frac{{Z_{c} - Z_{a}^{*} }}{{Z_{c} + Z_{a} }}$$

(2)

Γ:

Voltage reflection coefficient (0–1)

$\left| \varGamma \right|^{2}$ :

Power reflection coefficient (0–1)

V _in :

Input voltage at antenna terminals

where Z_c and Z_a are the impedances of chip and tag antennas

Fig. 2

Connection between antenna and chip

3 Proposed Tag Antenna Designs and Analysis

This section presents the design aspects of conventional planar radiating system that exhibits excellent impedance matching with commercially available RFID Chips. In the current scenario, the major requirement of general purpose RFID tags is the capability of reading even from distances greater than 1 m. It is evident from the literature survey that long range detection is possible with proper impedance matching, gain and threshold power.

It is a known fact that the radiating systems are resistive at resonant condition. It is also possible to enhance the reactance of the radiating system and accordingly many techniques are proposed. Among these are offset feed, parasitic patch, T-stub matching, probe stub matching (shorting pin), and curved surface patch etc., to name a few.

An attempt has been made to implement some of these techniques on the proposed hexagonal shape patch. The rectangle-shaped structure has a better efficiency and bandwidth characteristics which are better than circular disk even though the directivity is the same for both [10,11,12].

The aim of the antenna design is to match well with the frequently used though limited to a few commercial chips. The complex impedance of a few chips is tabulated and shown in Table 1 [4, 9, 13]. The chip impedance is almost constant throughout the band of frequencies [7]. The work is extended to design a tag antenna well matched with chip 1 of impedance 33 − j113 Ω [4, 13].

Table 1 Chip number and their impedance

The simulation results of an antenna are observed in terms of return loss (S₁₁), impedance plot, and radiation pattern (gain) at narrowband of operation 865.6–867.6 MHz. The designed antenna is thoroughly studied considering that these are bands recognized by India, America, Africa, Belgium, Denmark, etc. [14].

4 Design and Analysis of HPMSA

Moreover, the circular patch antenna has the advantage of less area occupancy than rectangular patch [10]. Better than the above two cases of rectangle and circular shape patches, incorporating the merits in both, a hexagonal structure is proposed here. The proposed structure area is 10% less than circular and 20% less than rectangular [10, 11]. An attempt has been made to achieve the required impedance to match the proposed structure.

4.1 Design of HPMSA

The proposed [10] HPMSA considers a patch of hexagonal shape on a RT-duroid substrate of high ε_r which is 10.0. Typical dimensions of the antenna substrate are 9 cm × 9 cm with a height of 0.156 cm simple probe feed (50 Ω). The dimension of the side arm of a hexagonal is calculated from basic circular patch dimensions given in [10], and the obtained value of ‘s’ is 3.516 cm at the desired operating frequency or 870 MHz. The simulation step is followed by fabrication-based validation in which the arm size is further optimized to 3.36 cm. The photograph of fabricated antenna top view (Patch) and bottom view (ground and feed) is shown in Fig. 3a, b.

Fig. 3

HPMSA designed. a Patch shape. b Ground plane (back side)

The return loss (Fig. 4) shows that antenna is resonated at 854 MHz with RL −4.978 dB. However, the patch performance is to be improved for the required application. The performance of antenna will be improved with slot technique and will be discussed in the next section.

Fig. 4

HPMSA return loss S₁₁ (dB) measured in network analyzer (NA)

4.2 Design of AHMSA

The HPMSA performance (Sect. 4.1) does not match the required frequency range. Hence, the technique of implementing a slot is considered and a similar hexagonal slot is made on the basic hexagonal patch. This is named as annular slot and hence the name AHMSA.

The merits of shells (i.e., slots) design are that it requires a smaller size than an orthodox hexagonal shape [11]. The slot models can be analyzed by using a cavity model [10].

4.3 Analysis of AHMSA

Let ‘a’ and ‘b’ be the inner and outer radius of an annular hexagonal and the resonance frequency can be given as [10, 11]

$$f_{\text{m}} = \frac{{K_{{nm}} c}}{{2\pi a\sqrt {\varepsilon_{\text{reff}} } }}$$

(3)

where k_nm are the roots of the characteristic equation

The modes are calculated from Eq. 4,

$$J_{n}^{'} \left( {kb} \right)Y_{n}^{'} \left( {ka} \right) - J_{n}^{'} \left( {ka} \right)Y_{n}^{'} \left( {kb} \right) = 0$$

(4)

where $Y_{n }^{'}$ and $J_{n}^{'}$ are derivative Bessel function of second and first kind of order n.

Equation 4 can be represented in another way as

$$J_{n}^{'} \left( {CX_{nm} } \right)Y_{n}^{'} \left( {X_{nm} } \right) - J_{n}^{'} \left( {X_{nm} } \right)Y_{n}^{'} \left( {CX_{nm} } \right) = 0$$

(5)

where C = b/a and

$$X_{nm} = K_{nm} a$$

(6)

$\varepsilon_{\text{r}}$ is replaced with effective permittivity $\varepsilon_{\text{e}}$. The $\varepsilon_{\text{e}}$ is given by Schneider [12] as

$$\varepsilon_{\text{e}} = \frac{1}{2}\left( {\varepsilon_{\text{r}} + 1} \right) + \frac{1}{2}\left( {\varepsilon_{\text{r}} - 1} \right)\left(1 + \frac{10t}{W}\right)^{{\frac{ - 1}{2}}}$$

(7)

where W = Microstrip of width = b − a.

According to the parallel plate model of circular ring, the microstrip line size is replaced with a parallel plate waveguide with identical $\varepsilon_{\text{re}}$ and Z_o. The modified inner and outer radii are

$$a_{\text{e}} = a - \frac{3h}{4}$$

(8)

$$b_{\text{e}} = b + \frac{3h}{4}$$

(9)

For the given values ‘a’ and ‘b’, effective a_e and b_e are calculated. Then the characteristic equation is solved by replacing ‘a’ and ‘b’ by a_e and b_e, respectively. After solving the characteristic equation for k_nm, the resonant frequency is determined from

$$f_{m} = \frac{{k_{nm} c}}{{2\pi \sqrt {\varepsilon_{\text{re}} } }}$$

(10)

• Characteristics of Proposed AHMSA:

The proposed antenna with an annular hexagonal ring is designed according to the rules specified in Sects. 4.2 and 4.3. The optimal design dimensions (from Eqs. 8 to 9) are finally obtained and simulated in the HFSS simulator (Fig. 5). The outer hexagon radius is 3.156 cm. The arm length corresponds to 3.09 cm. The inner hexagon is a slot etched at the center with a radius of 0.8 cm and arm length 0.69 cm (Fig. 6). The feed position is discussed below.

Fig. 5

Annular hexagonal microstrip antenna (AHMSA) in HFSS

Fig. 6

Fabricated antenna AHMSA, top view and ground plane

• Feed Position:

The output impedance of offset feed for the dominant mode at any distance from the center of the patch is given as [7]

$$R_{\text{in}} \left( {\rho^{\prime} = \rho_{\text{o}} } \right) = \frac{1}{{G_{\text{t}} }}\frac{{J_{1}^{2} \left( {K\rho_{\text{o}} } \right)}}{{J_{1}^{2} \left( {Ka_{\text{e}} } \right)}}$$

(11)

where $G_{\text{t}} \left( {\text{Total Conductance}} \right) = G_{\text{rad}} + G_{\text{c}} + G_{\text{d}}$ Radiation conductance

$$G_{\text{rad}} = \frac{{\left( {K_{0} a_{\text{e}} } \right)^{2} }}{480}\mathop \smallint \limits_{0}^{{\frac{\pi }{2}}} \left[ {J_{02}^{' 2} + \cos^{2} \theta J_{02}^{2} } \right]\sin \theta {\text{d}}\theta$$

(12)

Conduction due to ohmic loss

$$G_{\text{c}} = \frac{{\varepsilon_{\text{mo}} \pi \left( {\pi \mu_{\text{o}} f_{\text{r}} } \right)^{ - 3/2} }}{{4h^{2} \sqrt \sigma }}\left[ {\left( {Ka_{\text{e}} } \right)^{2} - m^{2} } \right]$$

(13)

Conduction due to dielectric loss

$$G_{\text{d}} = \frac{{\varepsilon_{\text{mo}} \tan \delta }}{{4\mu_{0} hf_{\text{r}} }}\left[ {\left( {ka_{\text{e}} } \right)^{2} - m^{2} } \right]$$

(14)

where a_e = effective radius.

Feed reactance

$$X_{f} = - \frac{\eta kh}{2\pi }\left[ {\ln \left( {\frac{kd}{4}} \right) + 0.577} \right]$$

(15)

where d is the diameter of the feed probe.

The maximum reactance is present when the probe is nearer to the corner, and it is reduced when the probe is moving toward the center.

The probe is connected at (−0.5 cm, 0, 0.156 cm) for better impedance matching called offset feeding [15]. Simulated and fabricated antennas are shown in Figs. 5 and 6, respectively.

The return loss (both simulated and measured) of AHMSA are shown in Fig. 7a, b. It is evident from the plot that the resultant geometry offers resonant frequency at 868.54 MHz (simulated) and 869.90 MHz (measured) at RL −10 dB. The bandwidth at 10 dB RL line is 865.30–871.30 MHz (simulated) and 868.42–871.64 MHz (measured), respectively. Simulated results match well with the measured results.

Fig. 7

Return loss of AHMSA. a Simulated. b Measured with NA. c Simulated impedance curve

The impedance versus frequency of AHMSA is shown in Fig. 7c. The impedance range at required band is 55.57 + j32.12 and 63.39 + j6.29 Ω. The RL (S₁₁ curve Fig. 7a) shows that the antenna is tuned at 868.54 MHz and 3-dB HP (half power) bandwidth range is 860.00–878.18 MHz.

The 3-dB HP range covers the required narrowband RFID tag operating band. The tag impedance (Z_tag) at 865.6 MHz is 55.56 + j32.12 and 63.38 + j6.29 Ω at 867.6 MHz.

• (iv) Performance of Different Slot Shapes

Till now, the performance of AHMSA with a hexagonal slot has been studied and analyzed. In order to arrive at the conclusion, it is required to analyze the performance with other different shaped slots in the HPMSA. For thin region, rectangular and pentagonal shaped patches are also considered and etched on HPMSA for analysis. The corresponding resonant frequencies and their bandwidth for rectangular, pentagonal, and hexagonal slots are shown in Table 2. The performance of hexagonal slot is matched to designed frequency and also nearer to the required band.

Table 2 Response of HPMSA with different slot shapes (simulated)

5 Impedance Matching Techniques

The impedance of an antenna at the measured band (Fig. 6) is not a perfect match to the selected chips. The following sections discuss the different techniques like parasitic patch, T-matching feed, and shorting pin to enhance the positive reactance of patch. The impedance values match the required IC chip 1. In each section, emphasis is laid on an analytical treatment of the technique followed by impedance matching measurements. Each technique mentioned in the following subsection is referred to different cases.

• (A) Parasitic Annular Hexa-MSA (PAHMSA)

In this method [4, 16, 17], a parasitic annular hexagonal structure is arranged around the actual annular hexagonal antenna. This parasitic element acts as an inductive element of the antenna (acts as reflector). The proposed antenna is shown in Fig. 8a. The top of the substrate is well used without increasing the dimension of the antenna.

Fig. 8

a PAHMSA simulated geometry in HFSS. b PAHMSA return loss (S₁₁) (simulated). c PAHMSA impedance plot (simulated)

The RL (S₁₁) plot Fig. 8b shows that the antenna is tuned at 874.87 MHz and 3-dB HP frequency range is 864.77–884.94 MHz. The 3-dB HP range covers the required RFID Band [14].

The impedance is noted from Fig. 8c at 865.6 MHz is 45.63 + j18.17 and 28.44 + 51.31 Ω at 867.6 MHz. The inductive nature of the impedance is increased and real value decreases more than in the previous case.

• (B) T-Stub Matching—AHMSA

The proposed antenna is shown in Fig. 9a. In this case, T-stub matching is proposed for better inductive matching of tag antenna to chip 1. It is a simple technique with no change in dimensions. The feed probe is located at the center of the patch and it is connected to the patch through T-stub matching as impedance converter [6, 18]. T-stub matching and their equivalent circuit are shown in Fig. 9b.

Fig. 9

T-stub matching AHMSA. a Proposed antenna. b T-matching (left), equivalent circuit (right)

From Fig. 9b, two types of microstrip lines have different impedances; Step in width exists at the junction. This type of discontinuity capacitance C_s represents the fringing field capacitance at the junction. An expression for C_s is given by

$$C_{\text{s}} \left( {\text{pF}} \right) = \sqrt {W_{1} W_{2} } \left[ {(10.1\log \left( {\varepsilon_{r} } \right) + 2.3)\frac{{W_{1} }}{{W_{2} }} - 12.6\log \varepsilon_{\text{r}} - 3.17} \right]$$

(16)

The inductances L₁ and L₂ are obtained from the total discontinuity inductance L_s as

$$L_{1} = \frac{{L_{W1} }}{{L_{W1} + L_{W2} }}L_{\text{s}}$$

(17)

$$L_{2} = \frac{{L_{W2} }}{{L_{W1} + L_{W2} }}L_{\text{s}}$$

(18)

where $L_{W}$ is inductance per unit length of microstrip of width W is given by

$$L_{W} = \frac{{Z_{0} \sqrt {\varepsilon_{\text{re}} } }}{c}$$

(19)

The expression for inductance $L_{\text{s}}$ is also given by

$$L_{\text{s}} \left( {\text{nH}} \right) = h\left[ {40.5\left( {\frac{{W_{1} }}{{W_{2} }} - 1.0} \right) - 75\log \frac{{W_{1} }}{{W_{2} }} + 0.2\left( {\frac{{W_{1} }}{{W_{2} }} - 1} \right)} \right]$$

(20)

(or)

$$L_{\text{s}} \left( {\text{nH}} \right) = 0.000987h\left( {1 - \frac{{Z_{01} }}{{Z_{02} }}} \right)^{2}$$

(21)

where h is the thickness of the substrate (µm).

Based on Eqs. 16–21, W₁ and W₂ are selected as 2 and 1 mm, respectively. The T-stub matching antenna (Fig. 9a) is analyzed. The return loss (S₁₁) curve (simulated) is shown in Fig. 10a, the antenna is tuned at 873.11 MHz and 3-dB HP frequency range is 865.30–881.81 MHz. The 3-dB HP range covers the required RFID band. The impedance curve is shown in Fig. 10b.

Fig. 10

a T-stub matching—AHMSA return loss plot (simulated). b T-Stub matching—AHMSA impedance plot (simulated)

The tag impedance (Z_tag) at 865.62 MHz is 41.18 + j85.04 and 64.83 + j76.95 Ω at 867.58 MHz.

• (C) T-Matching with Shorting Pin (Probe or Inductive Load) AHMSA

The inductive nature of an antenna is improved with selection of feed position, parasitic patch, and T-stub matching. It can be further improved with a shorting pin or shorting probe or an inductive loading technique. The shorting pin [19, 20] is connected between ground and patch through substrate.

The value of $\varepsilon_{\text{r}}$ is changed due to shorting pin and also it changes the resonant frequency [11]. One more advantage with shorting pin is that it reduces the effect on the results when placed on metallic objects. The post diameter is d (2 mm) and the strip width is W, for d/W ≪ 1. The pin offers the series and shunt inductances. The series inductance (L_S) is neglected and shunt inductance (L_P) taken on the circular wire is given as

$$L_{\text{P }} = 0.2h\left[ {\ln 4h/d + d/2h - 1} \right]$$

(22)

The series resistance (R) due to the loss in the wire is given in Eq. 23

$$R = 2.63*10^{ - 3} h/d\sqrt {\rho f/\rho C_{\text{u}} }$$

(23)

where $\rho$ is resistivity of the post metal and $\rho C_{\text{u}} = 1.72*10^{ - 4} \;\Omega$

The equivalent circuit of an antenna with a shorting pin is shown in Fig. 11. The shorting pin acts as parallel inductive load and it alters the current on the patch and affects the input impedance of the patch. The input impedance is calculated from Eq. 24. The input impedance of an antenna is

$$Z_{\text{in}} = \frac{{V_{\text{in}} }}{{I_{0} }}\;\Omega$$

(24)

where

V _in :

is the input voltage at feed point

I ₀ :

feed current

Fig. 11

Equivalent circuit of an antenna loading with shorting pin

The resultant input impedance of an antenna at any location on the patch is calculated by Eq. 25,

$$z_{\text{in}} = - j\omega \mu_{0} h\left\{ {\begin{array}{*{20}c} {\frac{1}{{\pi a^{2} k^{2} \left( {1 - j\delta } \right)}} + \mathop \sum \limits_{m = 2}^{\infty } \frac{{J_{0}^{2} \left( {k_{0m} \rho_{0} } \right)}}{{\pi a^{2} J_{0}^{2} \left( {k_{0m} a} \right)\left\{ {k^{2} \left( {1 - j\delta } \right) - k_{0m}^{2} } \right\}}}} \\ { + \frac{2}{\pi }\mathop \sum \limits_{m = 1}^{\infty } \mathop \sum \limits_{n = 1}^{\infty } \left( {\frac{\sin n\Delta }{n\Delta }} \right)^{2} \frac{{J_{n}^{2} \left( {k_{nm} \rho_{0} } \right)}}{{J_{0n}^{2} \left( {k_{nm} a} \right)}}\frac{{k_{0m}^{2} \cos^{2} n\emptyset_{0} }}{{\left\{ {k^{2} \left( {1 - j\delta_{\text{eff}} } \right) - k_{nm}^{2} } \right\}\left( {k_{nm}^{2} a^{2} - n^{2} } \right)}}} \\ \end{array} } \right\}$$

(25)

where $\delta_{\text{eff}}$ is called effective loss tangent of dielectric (including conductor loss, dielectric loss, and radiation loss).

When the shorting pin is placed at the corner of the patch, the normalized impedance (with 50 Ω) will increase w.r.to feed location when feed moves from edge of the patch to center and selects the best position of shorting pin. The proposed antenna in HFSS is shown in Fig. 12a and the designed antenna is shown in Fig. 12b, c. In this case, an inductive probe is added at an edge of the patch (green), along with T-stub (red) matching for better inductive matching of tag antenna to the chip 1.

Fig. 12

a T-matching with shorting pin AHMSA design (HFSS). b Designed antenna—top view. c Designed antenna—bottom view

It is a simple technique without altering the dimensions of an antenna. The antenna is analyzed and results are tabulated. Simulated return loss (Fig. 13a) shows that the antenna is tuned at 877.13 MHz and 3-dB HP range is 867.84–887.10 MHz. The simulated impedance response is shown in Fig. 13b. The tag impedances (Z_tag) at 865.6 and 867.6 MHz are 30.99 + j96.12, 49.08 + j101.57 Ω, respectively.

Fig. 13

Simulated. a Return loss. b Impedance plot. Measured with NA. c Return loss. d VSWR. e, f Radiation at frequencies at 865.62 and 867.63 MHz

The designed antenna is tested with NA and response of return loss (Fig. 13c) shows −13.84 dB at the band of interest even though resonated at out of the range at 674 MHz with RL −46 dB. The corresponding VSWR variation is shown in Fig. 2.14d. It shows that, the performance characteristic of the Simulated and measured are in good agreement.

The gain of the antenna is calculated at both ends of the observation band as shown in Fig. 13e, f. The gain is almost the same during the band, −3.43 dB. The beam is radiated along 0° and HPBW is 120° on φ = 0° plane.

The range performance of the antenna is calculated and tabulated in Tables 3 and 4. The range and PRC (power reflection coefficient) are calculated at chip threshold power −10 and −15 dBm at both band ends of frequencies 865.62 and 867.63 MHz. Tag impedance (Z_tag) matches to chip 1 more than the other chips. The minimum PRC at 865.62 MHz is 0.06 and read range is 5.78 m at −15 dBm threshold power. The maximum distance covered at 867.63 MHz is 5.81 m with a PRC 0.05 and the minimum distance with remaining chips is 3.18 m at 865.62 MHz.

Table 3 RFID tag read range measurement at frequency 865.62 MHz tag impedance (Z_tag) = 30.99 + j96.12 Ω, gain (G_tag) = −3.43 dB

Table 4 Read range measurement at frequency 867.6 MHz tag impedance (Z_tag) = 49.08 + j101.5 Ω, gain (G_tag) = −3.42 dB

• (D) Curved Shape Patch AHMSA

The proposed antenna is shown in Fig. 14a. In this case, the hexagonal patch edges are alternatively trimmed along with inductive probe and T-stub matching for better inductive matching of tag antenna to the chip and size reduction. It is a simple technique and the dimension of substrate is reduced from 9 to 7 cm (from 0.25λ to 0.2λ) length and width.

Fig. 14

a Simulated curved shape patch AHMSA. b Return loss. c Impedance matching

The return loss (S₁₁) shows (Fig. 14b) that the antenna is tuned at 877.89 MHz and 3-dB HP range is 867.84–888.63 MHz. The tag impedance (Z_tag) at 865.6 MHz is 26.75 + j86.88 Ω and 39.99 + j90.75 Ω at 867.6 MHz (Fig. 14c). This technique matches chip 1. The gain of the antenna is calculated at resonant frequency (f_r) (Fig. 2.15d)

• (E) Performance Comparison of Proposed Techniques

The response of each technique is compared with remaining techniques in terms of read range versus chip number at threshold power of −15 dBm at frequencies 865.6 and 867.6 MHz shown in Fig. 15a, b. The technique T-matching with probe (black line) is better than the remaining techniques with all chips.

Fig. 15

a, b Read range versus chip number at frequencies 865.6 and 867.6 MHz at P_th = −15 dBm

6 Conclusions

The system is designed for narrowband RFID Tag. The work started from HPMSA and achieved narrowband AHMSA. Its input impedance of antenna is well matched to chip 1. The minimum PRC 0.05 was achieved by introducing step-by-step inductive loading techniques to the basic HPMSA. The maximum read range 5.81 m with chip 1 and 3.64 m with chip 6 at −15 dBm threshold power was achieved. The proposed antenna (T-matching with stub AHMSA) was simple in design and analysis and well suited to the applications at selected frequency band.